Power factor controller

Abstract

A drive with a high impedance input, low impedance output is created. When a switching or driving action requiring the sourcing and sinking of current from a common node in a wide frequency range is desired, the invention allows the creation of a simple, efficient, two switch drive system that functions across a wide range of conditions. The circuit uses a discrete N-Channel FET paired with discrete PNP transistors. A high impedance input node is formed by connecting the FET gate to the transistor base. The differential threshold voltage that exists between the FET gate and the transistor base prevents the two devices from generating conflicting currents at the output node formed by the common source emitter. The circuit further lends itself to output waveform variations as may be required for various drive strategies by manipulating the input signal processing to custom modify the resulting output voltage and current.

Claims

We claim: 1. An AC to DC converter comprising: a first and second AC input line; an electromagnetic interference (EMI) filter circuit; an AC to DC rectifier circuit connected to the EMI filter circuit; a positive output terminal of the AC to DC rectifier circuit is connected to a first node of a primary winding of an inductor which has a magnetic element operating in a non-saturated region (NSME); a second node of the primary winding of the inductor is connected to an anode of a second rectifier and is connected to a drain of a N-Channel field effect transistor (FET); a source of the FET is connected to a negative node of the AC to DC rectifier circuit; and a controller circuit that modulates a FET conduction period to maintain a constant output voltage receives feedback from an output voltage of the converter. 2. The converter of claim 1 further comprising a bypass diode to conduct input transients around the converter to an output capacitor. 3. (FIG. 4) An AC to DC converter comprising: an AC input power connection; an EMI filter circuit connected to the AC power connection; a feedback circuit; a controller circuit; an inductor having a magnetic element operating in a non-saturated region (NSME); a line rectifier circuit; an output rectifier circuit; a switching circuit; said line rectifier circuit connected to an output of the EMI filter circuit; a positive output node of the line rectifier circuit connected to a first node of a primary winding of the inductor; a second node of the primary winding of the inductor is connected to an anode of the output rectifier circuit and to a drain of the switching circuit; a cathode of the output rectifier circuit is connected to a positive node of an output capacitor; a source of the FET is connected to a negative node of the line rectifier circuit; and the feedback circuit has a connection between a converter output and the controller, thereby providing a regulated DC output. 4. The converter of claim 3 further comprising a bypass diode to conduct input transients around the converter to an output capacitor. 5. An AC to DC power factor correcting boost converter, said boost converter comprising: an AC input power connection; an electromagnetic interference (EMI) filter circuit means functioning to block conducted EMI from the AC input power connection; an AC to DC rectifier circuit means functioning to convert an AC input voltage to a DC voltage output; said AC to DC rectifier circuit means connected between the EMI filter circuit means and a first node of a primary winding of a boost inductor means; said boost inductor means functioning to provide a boost voltage added to the DC voltage output using its magnetic element operating in a non-saturated region (NMSE); a second node of the primary winding of the boost inductor means is connected to an anode of a second rectifier and to an input node of an N-Channel switch means; said N-Channel switch means functioning to excite the primary winding of the boost inductor; an output node of the N-Channel switch means is connected to a negative node of the AC to DC rectifier circuit means; a controller circuit means functioning to modulate a trigger node of the N-Channel switch means; and a feedback circuit means connected between an output of the boost converter and the controller circuit means functioning to maintain a constant voltage output, thereby providing an EMI filtered, regulated, power factor corrected, non-isolated, DC line output. 6. The apparatus of claim 5 further comprising a bypass means functioning to conduct input transients directly around the boost converter to an output storage device means functioning to create an output from a transient through the storage drive means which is common to the boost converter output, thereby minimizing voltage drops while protecting the boost converter and its load.
CROSS REFERENCE APPLICATIONS This application is a continuation of application Ser. No. 09/923,027 filed Aug. 6, 2001 which is a divisional of application Ser. No. 09/410,849 filed Oct. 1, 1999 and issued as U.S. Pat. No. 6,272,025 on Aug. 7, 2001. FIELD OF INVENTION The present invention relates to converters, power supplies, more particularly, to single, or multi stage, AC/DC or DC/DC isolated and non-isolated push-pull converters including but not limited to, forward, flyback, buck, boost, push pull, and resonant mode converters, and power supplies, having individual or distributed NSME with high speed FET switching and efficient flyback management and or having input PFC (power factor correction) and input protection from lightning transients. The invention also allows the magnetic element(s) be distributed to accommodate packaging restrictions, multiple secondary windings, or operation at very high winding voltages. BACKGROUND OF THE INVENTION There are several basic topologies commonly used to implement switching converters. A DC-DC converter is a device that converts a DC voltage at one level to a DC voltage at another level. The converter typically includes a magnetic element having primary and secondary windings wound around it to form a transformer. By opening and closing the primary circuit at appropriate intervals control over the energy transfer between the windings occurs. The magnetic element provides an alternating voltage and current whose amplitude can be adjusted by changing the number and ratio of turns in each set of the windings. The magnetic element provides galvanic isolation between the input and the output of the converter. One of the topologies is the push-pull converter. The output signal is the output of an IC network that switches the transistors alternately “on” and “off”. High frequency square waves on the transistor output drive the magnetic element into AC (alternating current) bias. The isolated secondary outputs a wave that is rectified to produce DC (direct current). The push-pull converters generally have more components as compared to other topologies. The push-pull approach makes efficient use of the magnetic element by producing AC bias, but suffers from high parts count, thermal derating, oversized magnetics, and elaborate core reset schemes. The destructive fly-back voltages occurring across the switches are controlled through the use of dissipative snubber networks positioned across the primary switches. Another of the topologies is the forward converter. When the primary of the forward converter is energized, energy is immediately transferred to the secondary winding. In addition to the aforementioned issues the forward converter suffers from inefficient (dc bias) use of the magnetic element. The prior art power supplies use high permeability gapped ferrite magnetic elements. These are well known in the art and are widely used. The magnetics of the prior art power supplies are generally designed for twice the required power rating and require complex methods to reset and cool the magnetic elements resulting in increased costs and limited operating temperatures. This is because high permeability magnetic elements saturate during operation producing heat in the core, which increases permeability. and lowers the saturation threshold. This produces runaway heating, current spikes and/or large leakage currents in the air gap, reduced efficiency, and ultimately less power at higher temperatures and/or high load. The overall effects are, lower efficiency, lower power density, and forced air/heatsink dependant supplies that require over-rated ferrite magnetic elements for a given output over time, temperature, and loading. IMPROVEMENTS The combined improvements of the invention translate to higher system efficiencies, higher power densities, lower operating temperatures, and, improved thermal tolerance thereby reducing or eliminating the need for forced air cooling per unit output. The non-saturating magnetic properties are relatively insensitive to temperature (see FIG. 17 ), thus allowing the converter to operate over a greater temperature range. In practice, the operating temperature for the NSME is limited to 200C by wire/core insulation; the non-saturating magnetic material remains operable to near its Curie temperature of 500C. What are needed are converters having circuit strategies that make advantageous use of individual and distributed NSME. What are needed are converters having buffer circuits that provide fast, low impedance critically damped switching of the main FET's. What are needed are converters that incorporate efficient multiple “stress-less” flyback management techniques to rectify and critically damp excessive node voltages across converter switches. What are needed are converters having flux feedback frequency modulation. What are needed are converters that correct AC power factor. What is needed are converters that meet or exceed class B conducted EMI requirements. What are needed are converters tolerant of lightning and harsh thermal environments. The present invention addresses these and more. SUMMARY OF THE INVENTION The main aspect of the present invention is to implement converters having circuit strategies that make advantageous use of individual and distributed NSME for the achievement of the key performance enhancements disclosed herein. Another aspect of the present invention is to provide unique resonant tank circuit converter strategies with individual and distributed NSME that make use of higher primary circuit voltage excursions in the production of high frequency/high density magnetic flux. Another aspect of the present invention is a high energy density single stage frequency controlled resonant tank converter topology enabled by the use of individual and distributed NSME. Another aspect of the present invention is to provide a converter design that utilizes a FET drive technique consisting of an ultra fast, low RDS on N-channel FET for charging the main FET gate and an ultra fast P-channel transistor for discharging the main FET gate. Another aspect of the present invention is to provide converters that incorporate efficient multiple “stress-less” flyback management techniques to rectify and critically damp excessive node voltages across converter switches. Another aspect of the present invention is to provide a converter having core (flux) synchronized zero crossing frequency modulation. Another aspect of the present invention is to present a high power factor to the AC line. Another aspect of the present invention is to provide protection from high voltage (input line) transients. Another aspect of the present invention is to combine distributed magnetics advantageously with the other converter aspects. Another aspect of the present invention is active ripple rejection provided by several high-gain high-speed isolated control and feedback systems. Other aspects of this invention will appear from the following description and appended claims, reference being made to the accompanying drawings forming a part of this specification wherein like reference characters designate corresponding parts in the several views. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 and 1A is a schematic diagram of a two-stage power factor corrected AC to DC isolated output converter embodiment of the invention. FIG. 2 is a schematic diagram of a single stage DC to AC converter embodiment with isolated output sub-circuit DCAC 1 . FIG. 3 and 3A is a schematic diagram of a three stage AC to DC isolated output converter embodiment of the invention. FIG. 4 is a schematic diagram of a power factor corrected single stage AC to DC converter sub-circuit ACDFPF. FIG. 5 is a graph comparing typical winding currents in saturating and non-saturating magnetics of equal inductance. FIG. 6 is a schematic for a non-isolated low side switch buck converter sub-circuit NILBK. FIG. 7 is the preferred embodiment schematic for a tank coupled single stage converter sub-circuit TCSSC. FIG. 8 is a schematic for a tank coupled totem pole converter sub-circuit TCTP. FIG. 9 is a block diagram for a single stage non-isolated DC to DC boost converter NILSBST. FIG. 10 is a schematic for a two stage isolated DC to DC boost controlled push-pull converter BSTPP. FIG. 11 is a graph of permeability as a function of temperature for typical prior art magnetic element material. FIG. 12 is a graph of flux density as a function of temperature for typical prior art magnetic element material. FIG. 12A is a graph of magnetic element losses for various flux densities and operating frequencies typical of prior art magnetic element material. FIG. 13 is a graph showing standard switching losses. FIG. 14 is a graph showing lower switching losses of the invention. FIG. 15 is a graph showing the magnetizing curve (BH) for the NSME material. FIG. 16 is a graph of magnetic element losses for various flux densities and operating frequencies of the NSME material. FIG. 17 is a graph of permeability as a function of temperature for the NSME. FIG. 18 is a schematic representation of the boost NSME sub-circuit PFT 1 . FIG. 18A is a schematic representation of the NSME sub-circuit PFT 1 A. FIG. 18B is a schematic representation of the non-saturating two terminal NSME sub-circuit BL 1 . FIG. 18C is a schematic diagram of the NSME implemented as distributed magnetic assembly PFT 1 D. FIG. 19 is a schematic representation of the push-pull NSME sub-circuit PPT 1 . FIG. 19A is a schematic representation of the alternate push-pull NSME sub-circuit PPT 1 A. FIG. 20 is a schematic diagram of the NSME input transient protection and line filter sub-circuit LL. FIG. 21 is a schematic diagram of the alternate NSME input transient protection and line filter sub-circuit LLA. FIG. 22 is a schematic diagram of the AC line rectifier sub-circuit BR. FIG. 23 is a schematic diagram of the power factor controller sub-circuit PFA. FIG. 24 is a schematic diagram of the alternate power factor correcting boost control element sub-circuit PFB. FIG. 25 is a schematic diagram of the output rectifier and filter sub-circuit OUTA. FIG. 25A is a schematic diagram of an alternate rectifier sub-circuit OUTB. FIG. 25B is a schematic diagram of an alternate final output rectifier and filter sub-circuit OUTBB. FIG. 26 is a schematic diagram of the floating 18_Volt DC control power sub-circuit CP. FIG. 27 is a schematic diagram of the alternate floating 18_Volt DC push-pull control power sub-circuit CPA. FIG. 28 is a schematic diagram of the over temperature protection sub-circuit OTP. FIG. 29 is a schematic diagram of the high-speed low impedance buffer sub-circuit AMP, AMP 1 , AMP 2 and AMP 3 . FIG. 30 is a schematic diagram of the main switch snubber sub-circuit SN. FIG. 30A is a schematic diagram of the main switch rectifying diode snubber sub-circuit DSN. FIG. 31 is a schematic diagram of the alternate snubber sub-circuit SNA. FIG. 32 is a schematic diagram of the mirror snubber sub-circuit SNB. FIG. 33 is a schematic diagram of the pulse-width/Frequency modulator sub-circuit PWFM. FIG. 34 is an oscillograph of node voltages measured during operation of sub-circuit PWFM (FIG. 33 ). FIG. 35 is an oscillograph of the primary tank voltage measured during operation of sub-circuit TCTP (FIG. 8 ). FIG. 36 is a schematic diagram of the non-isolated18-Volt DC control power sub-circuit REG. FIG. 37 is a schematic for a non-isolated high-side switch buck converter sub-circuit HSBK. FIG. 38 is a schematic for the low-side buck regulated two-stage converter embodiment with isolated push-pull output sub-circuit LSBKPP. FIG. 39 is a schematic for an alternate isolated two-stage low-side switch buck converter sub-circuit LSBKPPBR. FIG. 40 is a schematic diagram of the over voltage feed back sub-circuit IPFFB. FIG. 40A is a schematic diagram of the non-isolated boost output voltage feedback sub-circuit FBA. FIG. 40B is a schematic diagram of the isolated output voltage feedback sub-circuit IFB. FIG. 40C is a schematic diagram of the alternate isolated over voltage feedback sub-circuit IOVFB. FIG. 41 is a schematic diagram of the non-isolated output voltage feedback sub-circuit FBI. FIG. 42 is a schematic diagram of an over voltage protection sub-circuit OVP. FIG. 42A is a schematic diagram of the isolated over voltage feedback sub-circuit OVP 1 . FIG. 42B is a schematic diagram of the over voltage protection sub-circuit OVP 2 . FIG. 42C is a schematic diagram of the isolated over voltage feedback sub-circuit OVP 3 . FIG. 43 is a schematic diagram of the Push-pull oscillator sub-circuit PPG. Before explaining the disclosed embodiments of the present invention in detail, it is to be understood that: The invention is not limited in its application to the details of the particular arrangements shown or described, since the invention is capable of other embodiments. The expression “distributed magnetic(s)” refers to the configuration of multiple magnetic elements that share a single series coupled primary winding to induce isolated output currents from multiple series or parallel secondary windings. Also, the terminology used herein is for the purpose of description not limitation. DESCRIPTION OF THE PREFERRED EMBODIMENT In this and other descriptions contained herein, the following symbols shall have the meanings attributed to them: “+” shall indicate a series connection, such as resistor A in series with resistor B shown as “A+B”. “||” Shall indicate a parallel connection, such as resistor A in parallel with resistor B shown as “A||B”. Referring first to FIG. 7, a schematic diagram of the preferred embodiment of the invention. FIG. 7 is a schematic of the preferred embodiment of a tank coupled single stage converter sub-circuit TCSSC. Sub-circuit TCSSC consists of resistor R 20 and RLOAD, capacitor C 10 , transistors Q 21 and Q 11 , sub-circuit CP (FIG. 26 ), sub-circuit PFT 1 (FIG. 18 ), sub-circuit OUTA (FIG. 25 ), sub-circuit AMP (FIG. 29 ), sub-circuit IFB (FIG. 40B) and sub-circuit PWFM (FIG. 33 ). FIG. Table 7 Value/part Element number R20 1k ohms R61 2k ohms Q21 TST541 U12 4N29 Q11 IRFP460 C10 1.8 uf CSSC can be configured to operate as an AC-DC converter, a C-DC converter, a DC-AC converter, and an AC-AC converter. Sub-circuit TCSSC consists of resistor R 20 and RLOAD, capacitor C 10 , switches Q 11 and Q 21 , opto-isolator U 12 , sub-circuit PFT 1 (FIG. 18 ), sub-circuit OUTA (FIG. 25 ), sub-circuit CP (FIG. 26 ), sub-circuit AMP (FIG. 29 ), sub-circuit IFB (FIG. 40B) and sub-circuit PWFM (FIG. 33 ). External power source VBAT connects to pins DCIN+ and DCIN−. Source power may also be derived from rectified AC line voltage such as FIG. 20 or FIG. 21 to form a single stage power factor corrected AC to DC converter with isolated output. From DCIN+ resistor R 20 connects to sub-circuit CP pin CP+, sub-circuit AMP pin GA+, U 12 LED anode and to sub-circuit PWFM pin PWFM+. Resistor R 20 provides startup power to the converter until the control supply regulator sub-circuit CP reaches the desired 18 -volt output. VBAT negative is the ground return node connects to sub-circuit PWFM pin PWFM 0 , Q 11 source, sub-circuit AMP pin GA 0 , sub-circuit CP pin CT 0 , pin DCIN− and sub-circuit PFT 1 pin S 1 CT. Magnetic element winding node S 1 H of sub-circuit PFT 1 is connected to CP pin CT 1 A. Magnetic element winding node S 1 L of sub-circuit PFT 1 is connected to CP pin CT 2 A. Sub-circuit PWFM is designed as a constant 50% duty-cycle variable frequency generator. Sub-circuit PWFM Clock output pin CLK is connected to input of buffer sub-circuit AMP pin GA 1 . The output of buffer sub-circuit AMP pin GA 2 is connected to the gate of Q 11 and R 21 . Resistor R 21 is connected to the cathode of U 12 LED. The emitter of Q 21 and drain of Q 11 is connected to sub-circuit PFT 1 pin P 1 A. Pin P 1 B of sub-circuit PFT 1 is connected through tank capacitor C 10 to node DCIN+, Q 21 collector and through resistor R 61 to U 12 phototransistor collector. The emitter of U 12 phototransistor is connected to the base of Q 21 . With PWFM pin CLK high transistor Q 11 conducts charging capacitor C 10 through NSME PFT 1 from VBAT storing energy in PFT 1 . Sub-circuit PWFM switches CLK low, Q 11 turns “off”. With CLK low LED of U 12 is turned “on” injecting base current into Q 21 . With transistor Q 21 “on” the tank circuit is completed, allowing capacitor C 10 to discharge into NSME PFT 1 winding 100 (FIG. 18 ). Now the energy not transferred into the load is released from NSME PFT 1 into the now forward biased NPN switch Q 21 back into capacitor C 10 . Thus any energy not used by the secondary load remains in the tank coupled primary circuit (winding 100 ). When the switching occurs at the resonant frequency, high voltages oscillate between C 10 and winding 100 creating high flux density AC excursions in PFT 1 . C 10 and PFT 1 exchange variable AC currents whose magnitude is controlled by frequency modulation scheme IFB and PWFM. The large primary voltage generates large, high frequency biases in the NSME PFT 1 thereby producing high flux density AC excursions to be harvested by secondary windings 102 and 103 (FIG. 18) to support a load or rectifier sub-circuit OUTA. Magnetic element winding node S 2 H of sub-circuit PFT 1 is connected to OUTA pin C 7 B. Magnetic element winding node S 2 L of sub-circuit PFT 1 is connected to OUTA C 8 B. Magnetic element winding node S 2 CT of sub-circuit PFT 1 is connected to OUTA pin OUT−. Node OUT− is connected to RLOAD, pin B− and to sub-circuit IFB pin OUT−. Rectified power is delivered to pin OUT+ of OUTA and is connected to RLOAD, pin B+ and to sub-circuit IFB pin OUT+. Sub-circuit IFB provides the isolated feedback signal to the sub-circuit PWFM. Frequency control pin FM 1 of sub-circuit PWFM is connected to sub-circuit IFB pin FBE. Internal reference pin REF of sub-circuit PWFM is connected to sub-circuit IFB pin FBC. PWFM is designed to operate at the resonate frequency of the tank (2*pi*(square root (C 10 *inductance of 100 (FIG. 18 )). When sub-circuit IFB senses the converter output is at the target voltage, current from PWFM pin REF is injected into FM 1 . Injecting current into FM 1 commands the PWFM to a lower clock frequency pin CLK. Driving the tank out of resonance reduces the amount of energy added to the tank thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM off or 0 Hz, i.e.: at no load, all primary activity stops. The input current from VBAT may be steady state or variable DC. When TCSSC is operated from rectified AC (sub-circuit LL FIG. 20 ), high input (line) power factor and input transient protection is achieved. The primary and secondary currents of PFT 1 are sinusoidal and free of edge transitions making the converter very quiet. In addition the switches Q 11 and Q 21 are never exposed to the large circulating voltage induced in the tank (See FIG. 35 ). This allows the use of lower voltage switches in the design thereby reducing losses and increasing the MTBF. Sub-circuit TCSSC takes advantage of the desirable properties of the NSME in this converter topology. TCSSC is well suited for implementation with distributed NSME PFT 1 D (FIG. 18 C). This combination exemplifies how distributed magnetics enable advantageous high voltage converter design variations that support form factor flexibility and multiple parallel secondary outputs from series coupled voltage divided primary windings across multiple NSME. This magnetic strategy is useful in addressing wire/core insulation, form factor and packaging limitations, circuit complexity and manufacturability. These converter strategies are very useful for obtaining isolated high current density output from a high voltage low current series coupled primary. Adjusting the secondary turn's ratio allows TCSSC to generate very large AC or DC output voltages as well as low-voltage high current outputs. ADDITIONAL EMBODIMENTS FIG. 1 and 1A is a schematic diagram of a two stage power factor corrected AC to DC converter. The invention is comprised of line protection filter sub-circuit LL (FIG. 20) and full-wave rectifier sub-circuit BR (FIG. 22 ). A power factor corrected regulated boost stage with sub-circuits PFA 2 (FIG. 23 ), snubber sub-circuit SN (FIG. 30 ), magnetic element sub-circuit PFT 1 (FIG. 18 ), sub-circuit CP (FIG. 26 ), buffer sub-circuit AMP (FIG. 29 ), over temperature sub-circuit OTP (FIG. 28 ), over voltage feedback sub-circuit IPFFB (FIG. 40) and voltage feedback sub-circuit IFB (FIG. 40 B). Start up resistor R 2 , filter capacitor C 1 , PFC capacitor C 2 , flyback diode D 4 , switch transistor Q 1 , hold up capacitors C 17 and C 16 , and resistor R 17 . An efficient push-pull isolation stage with sub-circuits CPA (FIG. 27 ), PPG (FIG. 43 ), AMP 1 (FIG. 29 ), AMP 2 (FIG. 29 ), snubber sub-circuits SNB (FIG. 32) and SNA (FIG. 31 ), resistor Rload, transistors Q 6 and Q 9 , magnetic element PPT 1 (FIG. 19 ), and OUTA (FIG. 25 ). Table FIG. 1 Value/part Element number C1 0.01 uf C2 1.8 uf R2 100k ohms D4 8A,600 V Q1 IRFP 460 C17 100 uf C16 100 uf R17 375k ohms Q6 FS 14SM-18A Q9 FS 14SM-18A In the two-stage converter the primary side voltage to the second push-pull output stage is modulated by the power factor corrected input (boost) stage. Each stage can comprise of individual and distributed NSME. A graph of B-H hysteresis for the non-saturating magnetics is set forth in FIG. 15 . Although the following description is in terms of particular converter topologies, i.e., flyback controlled primary and constant duty cycle push pull secondary, number of outputs, the style, and arrangement of the several topologies are offered by way of example, not limitation. In addition non-saturating magnetics BL 1 , PFT 1 , and PPT 1 may be implemented as distributed NSME. As an example PFT 1 is shown as a distributed magnetic PFT 1 A (FIG. 18 C). Distributed magnetics enable advantageous high voltage converter design variations that support form factor flexibility and multiple parallel secondary outputs from series coupled voltage divided primary windings across multiple NSME. The negative of the hold-up capacitor(s) [C 17 ||C 16 ] is connected to bridge positive. This allows the rectified line voltage to be excluded from the boost voltage in the hold-up capacitor(s). This, in turn, allows direct regulation of the push-pull stage from the boost (PFC) stage. This eliminates the typical PWM control of the oversized thermally derated transformer and many sub-circuit components from the known art. AC line is connected to sub-circuit LL (FIG. 20) between pins LL 1 and LL 2 . AC/earth ground is connected to node LL 0 . The filtered and voltage limited AC line appears on node/pin LL 5 of sub-circuit LL and connected to node BR 1 of bridge rectifier sub-circuit BR (FIG. 22 ). The neutral/AC return leg of the filtered and voltage limited AC appears on pin LL 6 of sub-circuit LL is connected to input pin BR 2 of BR. The line voltage is full-wave rectified and is converted to a positive haversine appearing on node BR+ of sub-circuit BR (FIG. 22 ). Start up resistor R 2 connects BR+ to sub-circuit CP pin CP+. Node CP+ connects to pins PFA+ of control element sub-circuit PFA (FIG. 23) and over temperature switch sub-circuit OTP (FIG. 28) pin GAP. Resistor R 2 provides start up power to the control element until the rectifier/regulator CP is at full output. Node S 1 H from PFT 1 is connected to node PFVC of sub-circuit PFA. The zero crossings of the core are sensed when the voltage at S 1 H is at zero. The core zero crossings are used to reset the PFC and start a new cycle. The positive node of the DC side of bridge BR+ is connected through capacitor C 2 to BR−. C 2 is selected for various line and load conditions to de-couple switching current from the line improving power factor while reducing line harmonics and EMI. Primary of NSME sub-circuit PFT 1 (FIG. 18) pins P 1 B and S 2 CT connects to pin SNL 1 of snubber sub-circuit SN (FIG. 30 ), to sub-circuit BR pin BR+ and connects to pin BR+ (FIG. 1 A). The return line for the rectified AC power BR− is connected to the following pins: BR− of sub-circuit BR, PFA pin BR−, sub-circuit AMP pin GA 0 , output switch Q 1 source, capacitor C 2 , sub-circuit CP pin CT 0 sub-circuit PFT 1 pin S 1 CT and CT 20 through EMI filter capacitor C 1 to earth ground node LL 0 . Pin BR+ from FIG. 1 is connected to FIG. 1A sub-circuits CPA pin SN pin SNL 1 , sub-circuit PFT 1 pin P 1 B, and sub-circuit PFT 1 pin S 2 CT. Pin BR+ continues to FIG. 1A connecting to sub-circuit CPA pin CT 20 , PPG (FIG. 43) pin PPG 0 , sub-circuit AMP 1 pin GA 0 , sub-circuit AMP 2 pin GA 0 , sub-circuit IPFFB pin PF−, Capacitor [C 16 ||C 17 || resistor R 17 ], transistor Q 6 source, transistor Q 9 source, sub-circuit SNA pin SNA 2 and sub-circuit SNB pin SNB 2 . The drain of output switch Q 1 is connected to diode D 4 anode, sub-circuit SNB pin SNL 2 , and sub-circuit PFT 1 pin P 1 A and sub-circuit SN pin SNL 2 . Snubber network SN reduces the high voltage stress to Q 1 until flyback diode D 4 begins conduction. Line coupled, power factor corrected boost regulated output voltage of the AC to DC converter stage (FIG. 1) appears on node PF+. Addition efficiency may be realized by connecting sub-circuit DSN (FIG. 30A) in parallel with D 4 . The regulated boost output PF+ connects to the following: sub-circuit SN pin SNOUT, sub-circuit DSN pin SNOUT and diode D 4 cathode. Node PF+ also connects on FIG. 1A to capacitors [C 16 ||C 17 ||R 17 ], sub-circuit. IPFFB 15 (FIG. 40) pin PF+, sub-circuit PPT 1 (FIG. 19) pin P 2 CT, snubber sub-circuit SNA (FIG. 31) pin SNA 3 , and snubber SNB (FIG. 32) pin SNB 3 . Magnetic element winding pin S 1 H of sub-circuit PFT 1 is connected to CP pin CT 1 A and pin PFVC of sub-circuit PFA. Magnetic element winding node S 1 L of sub-circuit PFT 1 is connected to CP pin CT 2 A. Magnetic element winding node S 2 H of sub-circuit PFT 1 is connected to pin 10 FIG. 1A then to CPA pin CT 1 B. Magnetic element winding node S 2 L of sub-circuit PFT 1 is connected to pin 12 FIG. 1A then to CPA pin CT 2 B. Sub-circuit PFA using the AC line phase, load voltage, and magnetic element feedback, generates a command pulse PFCLK. Pin PFCLK of sub-circuit PFA (FIG. 23) is connected to the input of buffer amplifier pin GA 1 of sub-circuit AMP 1 (FIG. 29 ). Buffered high-speed gate drive output pin GA 2 of sub-circuit AMP is connected to gate of switch FET Q 1 . The buffering provided by AMP shortens switch Q 1 ON and OFF times greatly reducing switch losses (see FIG. 13 & 14 ). The source of Q 1 with pin GA 0 is connected to return node BR−. Power to sub-circuit AMP is connected to pin GA+ from sub-circuit OTP pin TS+. Thermal switch THS 1 is connected to Q 1 . In the event the case of Q 1 reaches approximately 105 C THS 1 opens removing power to sub-circuit AMP, safely shutting down the first (input) stage. Normal operation resumes after the switch temperature drops 20-30 deg. C closing THS 1 . Drain of output switch Q 1 is connected to primary winding pin P 1 A of non-saturating magnetic sub-circuit PFT 1 (FIG. 18) and to pin SNL 2 of snubber sub-circuit SN (FIG. 30 ). Reference voltage from PFC sub-circuit PFA pin PFA 2 is connected to feedback networks sub-circuit IPFFB pin FBC and to sub-circuit IFB pin FBC. Control current feedback networks is summed at node PF 1 of sub-circuit PFA. Pin PF 1 is connected to feed back networks sub-circuit IPFFB pin FBE and to sub-circuit IFB pin FBE. Constant frequency/duty-cycle non-overlapping two-phase generator sub-circuit PPG (FIG. 43 1 A) generates the drive for the push-pull output stage. Phase one output pin PHI is connected to sub-circuit AMP 1 pin GA 1 , second phase output pin PH 2 is connected to sub-circuit AMP 2 pin GA 1 . Output of amplifier buffer sub-circuit AMP 1 pin GAP 2 connects to gate of push-pull output switch Q 6 . Output of amplifier buffer sub-circuit AMP 2 pin GAP 2 connects to gate of push-pull output switch Q 9 . The buffering currents from AMP 1 and AMP 2 provide fast, low impedance critically damped switching to Q 6 and Q 9 greatly reducing ON-OFF transition time and switching losses. Regulated 18-volt power from sub-circuit CPA (FIG. 1A) pin CP 2 + is connected to amplifier buffer sub-circuit AMP 1 pin GA+, amplifier buffer sub-circuit AMP 2 pin GA+ and sub-circuit PPG pin PPG+. Drain of transistor Q 6 is connected to snubber network sub-circuit SNB pin SNB 1 and to non-saturating center tapped primary magnetic element sub-circuit PPT 1 pin P 2 H. Drain of transistor Q 9 is connected to snubber network sub-circuit SNA (FIG. 31) pin SNA 1 and sub-circuit PPT 1 pin P 2 L. Source of transistor Q 6 is connected to snubber network sub-circuit SNB pin SNB 2 , transistor Q 9 source, sub-circuit SNA pin SNA 2 and to return node BR+. Isolated output of NSME sub-circuit PPT 1 pin SH connects to Pin C 7 B of rectifier sub-circuit OUTA (FIG. 25 A), pin SL connects to sub-circuit OUTA C 8 B. Center tap of PPT 1 pin SCT is the output return or negative node OUT− it connects to sub-circuit OUTA pin OUT− and sub-circuit IFB (FIG. 40B) pin OUT− and RLOAD. Converter positive output from sub-circuit OUTA pin OUT+ is connected to RLOAD and sub-circuit IFB pin OUT+. FIG. 1 elements LL 1 , BR, PFA, AMP, Q 1 , IPFFB, IFB and PFT 1 (input stage) perform power factor corrected AC to DC conversion. The regulated high voltage output of this converter supplies the efficient fixed frequency/duty-cycle push-pull stage comprising PPG, AMP 1 , AMP 2 , Q 6 , Q 9 , PPT 1 and OUTA (FIG. 1 A). Magnetic element sub-circuit PPT 1 provides galvanic isolation and minimal voltage overshoot and ripple in the secondary thus minimizing filtering requirements of the rectifier sub-circuit OUTA. Five volt reference output from sub-circuit PFA pin PFA 2 connects to pin 15 then to FIG. 1A sub-circuit IPFFB pin FBC and to sub-circuit IFB pin FBC. Pulse width control input from sub-circuit PFA pin PF 1 connects to pin 14 then to FIG. 1A sub-circuit IPFFB pin FBE and to sub-circuit IFB pin FBE. Sub-circuit IFB provides high-speed feedback to the AC DC converter, the speed of the boost stage provides precise output voltage regulation and active ripple rejection. In the event of sudden line or load changes, sub-circuit IPFFB corrects the internal boost to maintain regulation at the isolated output. Remote load sensing and other feedback schemes known in the art may be implemented with sub-circuit IPFFB. This configuration provides power factor corrected input transient protection, rapid line-load response, excellent regulation, isolated output and quiet efficient operation at high temperatures. FIG. 2 is a schematic diagram of an embodiment of a DC to AC converter. The invention DCAC 1 is an efficient push-pull converter. Comprised of sub-circuits PPG (FIG. 43 ), AMP 1 (FIG. 29 ), AMP 2 (FIG. 29 ), SNB (FIG. 32 ), SNA (FIG. 31 ), PPT 1 (FIG. 19) and OUTA (FIG. 25 ), switches Q 6 and Q 9 . FIG. 2 Table Element Value/part number Q6 FS 14SM-18A Q9 FS 14SM-18A Converter ACDC 1 accepts variable DC voltage and efficiently converts it to a variable AC voltage output at a fixed frequency. Variable frequency operation may be achieved by simple changes to PPG. In this embodiment fixed frequency operation is required. The magnetic element comprises non-saturating magnetics. A graph of B-H hysteresis for the non-saturating magnetics is set forth in FIG. 15 . Variable DC voltage is applied to pin DC+. The pin DC+ connects to the following, sub-circuit PPT 1 (FIG. 19) pin P 2 CT, snubber sub-circuit SNA (FIG. 31) pin SNA 3 , and snubber SNB (FIG. 32) pin SNB 3 . Constant frequency non-overlapping two-phase generator sub-circuit PPG (FIG. 43) generates the drive for the push-pull output switches. Phase one output pin PH 1 is connected to sub-circuit AMP 1 pin GA 1 , the second phase output pin PH 2 is connected to sub-circuit AMP 2 pin GA 1 . Output of amplifier buffer sub-circuit AMP 1 pin GAP 2 connects to gate of push-pull output switch Q 6 . Output of amplifier buffer sub-circuit AMP 2 pin GAP 2 connects to gate of push-pull output switch Q 9 . The buffering provided by AMP 1 and AMP 2 shortens switch Q 1 ON and OFF times greatly reducing switching losses (See FIG. 13 and 14 ). External regulated 18-volt power from pin P 18 V connected to amplifier buffer sub-circuit AMP 1 pin GA+, amplifier buffer sub-circuit AMP 2 pin GA+ and sub-circuit PPG pin PPG+. Drain of transistor Q 6 is connected to snubber network sub-circuit SNB pin SNB 1 and to non-saturating center tapped primary magnetic element sub-circuit PPT 1 pin P 2 H. Drain of transistor Q 9 is connected to snubber network sub-circuit SNA (FIG. 31) pin SNA 1 and sub-circuit PPT 1 pin P 2 L. Source of transistor Q 6 is connected to snubber network sub-circuit SNB pin SNB 2 , transistor Q 9 source, sub-circuit SNA pin SNA 2 , sub-circuit AMP 1 pin GA 0 , sub-circuit AMP 2 pin GA 0 , sub-circuit PPG pin PPG 0 , and to return pin DC−. AC output of NSME sub-circuit PPT 1 pin SH connects to Pin ACH, pin SL connects to pin ACL. Center tap of PPT 1 pin SCT is connected to pin AC 0 . Magnetic element sub-circuit PPT 1 provides galvanic isolation and minimal voltage overshoot in the secondary thus minimizing filtering requirements if a rectifier assembly is attached. Sub-circuit DCAC 1 may be used as a stand-alone converter or as a fast quiet efficient stage in a multi stage converter system. Sub-circuit DCAC 1 achieves isolated output, quiet operation, efficient conversion, and operation at. high and low temperatures. FIG. 3 and 3A is a three-stage version of the present invention. The arrangement is comprised of an AC-DC or DC-DC boost converter stage, DC-DC forward converter. stage, and a push-pull stage. This system reduces losses by combining low current buck regulation, buffered switching, rectified snubbering, and NSME in each stage. A power factor corrected boost stage is used to assure that any load connected to the converter looks like a resistive load to the AC line, eliminating undesirable harmonic and displacement currents in the AC power line. NSME having a lower permeability compared to the prior art are used to minimize magnetizing losses, improve coupling efficiency, minimize magnetic element heating, eliminate saturated core current spikes/gap leakage, reduce parts count, reduce thermal deterioration, and increase MTBF (mean time before failure). The invention also uses an emitter follower circuit with a high speed switching FET to slew the main FET gate rapidly. The use of non-saturating magnetics allows operation at higher voltages, which proportionally lowers current further reducing switch, magnetic element, and conductor losses due to I 2 R heating. High voltage FET switches have the added benefit of lower gate capacitance, which translates to faster switching. At turn on, the n-channel gate drive FET quickly charges the main FET gate. At turn off, a PNP Darlington transistor switch quickly discharges the main FET gate. The flyback effect in the PFC stage is managed by use of rectifying RC networks positioned across the output diode with an additional capacitor coupled diode across the switched magnetic element to decouple and further dampen the inductive flyback. FIG. 3 and FIG. 3A is a schematic diagram of a three stage AC to DC converter. FIG. 3 and 3A is a three-stage version of the present invention. The arrangement is comprised of an AC-DC or DC-DC boost converter stage, DC-DC forward converter stage, and a push-pull stage. This system reduces losses by combining low current buck regulation, buffered switching, rectified snubbering, and NSME in each stage. A power factor corrected boost stage is used to assure that any load connected to the converter looks like a resistive load to the AC line, eliminating undesirable harmonic and displacement currents in the AC power line. NSME having a lower permeability compared to the prior art are used to minimize magnetizing losses, improve coupling efficiency, minimize magnetic element heating, eliminate saturated core current spikes/gap leakage, reduce parts count, reduce thermal deterioration, and increase MTBF (mean time before failure). The invention also uses an emitter follower circuit with a high speed switching FET to slew the main FET gate rapidly. The use of non-saturating magnetics allows operation at higher voltages, which proportionally lowers current further reducing switch, magnetic element, and conductor losses due to I 2 R heating. High voltage FET switches have the added benefit of lower gate capacitance, which translates to faster switching. At turn on, the n-channel gate drive FET quickly charges the main FET gate. At turn off, a PNP Darlington transistor switch quickly discharges the main FET gate. The flyback effect in the PFC stage is managed by use of rectifying RC networks positioned across the output diode with an additional capacitor coupled diode across the switched magnetic element to decouple and further dampen the inductive flyback. The invention is comprised of a power factor corrected regulating boost stage with line protection filter sub-circuit LL 1 (FIG. 21) and full-wave rectifier sub-circuit BR (FIG. 22) and capacitors C 1 and C 2 . Sub-circuits PFB (FIG. 24 ), resistor R 2 , rectifier CP (FIG. 26 ), magnetic element PFT 1 (FIG. 18 ), over temperature protection OTP (FIG. 28) snubber SN (FIG. 30) gate buffer AMP (FIG. 29 ), switch transistors Q 1 , flyback diode D 4 , holdup capacitors C 17 and C 16 , bleed resistor R 17 , and voltage feedback sub-circuit FBA (FIG. 40 A). An efficient second pre-regulating buck stage with sub-circuits PWFM (FIG. 33 ), current sense resistor R 26 , rectifier CPA (FIG. 27 ), magnetic element BL 1 (FIG. 18 B), over voltage protection OVP (FIG. 42 ), IPFFB (FIG. 40) gate buffer AMP 3 (FIG. 29 ), switch transistor Q 2 , flyback diode D 70 , storage capacitor C 4 , and voltage feedback sub-circuit IFB (FIG. 40 B). An efficient third push-pull isolation stage with sub-circuits CPA (FIG. 27 ), two-phase generator PPG (FIG. 43 ), gate buffers AMP 1 (FIG. 29) and AMP 2 (FIG. 29 ), switch transistors Q 6 , and Q 9 , snubbers SNA (FIG. 31) and SNB (FIG. 32 ), magnetic element PPT 1 (FIG. 19) and rectifier OUTA (FIG. 25 ). FIG. Table 3 3A Value/part Element number C1 .01 uf C2 1.8 uf R2 100k ohms D4 STA1206 DI R17 375k ohms Q1 IRFP460 C16 100 uf C17 100 uf R26 .05 ohms D70 STA1206 DI Q2 IRFP460 C4 10 uf Q6 FS14Sm-18A Q9 FS14Sm-18A AC line is connected to sub-circuit LLA (FIG. 21) between pins LL 1 and LL 2 . AC/earth ground is connected to node LL 0 . The filtered and voltage limited AC line appears on node/pin LL 5 of sub-circuit LLA and connected to node BR 1 of bridge rectifier sub-circuit BR. The neutral/AC return leg of the filtered and voltage limited AC appears on pin LL 6 of sub-circuit LL is connected to input pin BR 2 of BR. The line voltage is full-wave rectified and is converted to a positive haversine appearing on node BR+ of sub-circuit BR. Start up resistor R 2 connects BR+ to sub-circuit CP pin CP+. Node CP+ connects to pins PFA+ of control element sub-circuit PFB and over temperature switch sub-circuit OTP pin GAP. Resistor R 2 provides start up power to the control element until the regulator CP is at full output. Node S 1 H from PFT 1 is connected to pin 31 (FIG. 3) then to pin CT 1 A of sub-circuit CP and pin PFVC of sub-circuit PFB. The zero crossing of the core bias are sensed when the voltage at S 1 H is at zero relative to BR−. The core zero crossings are used to reset the PFC and start a new cycle. The positive node of the DC side of bridge BR+ is connected through capacitor C 2 to BR−. Capacitor C 2 is selected for various line and load conditions to de-couple switching current from the line improving power factor. Sub-circuit BR pin BR+ connects to pin SNL 1 of snubber sub-circuit SN, sub-circuit PFB pin BR+ and pin BR+ (FIG. 3A) then to primary of NSME sub-circuit PFT 1 pin P 1 B and to sub-circuit OVP pin BR+. The return line for the rectified AC power is connected to the following pins; BR− of sub-circuit BR, sub-circuit PFT 1 pin S 1 CT, PFC sub-circuit PFB pin BR−, sub-circuit FBA pin BR−, capacitor C 2 , sub-circuit CP pin CT 0 , sub-circuit IPFFB pin FBE, and through EMI filter capacitor C 1 to earth ground node LL 0 . Node BR− continues to FIG. 3A connecting to R 26 , capacitors [C 16 ||C 17 ||R 17 ], sub-circuit OVP pin BR−, sub-circuit PWFM pin PWFM 0 , sub-circuit AMP 3 pin GA 0 , switch Q 2 source. Floating ground node PF− is connected to magnetic element sub-circuit PFT 1 pin S 2 CT, rectifier sub-circuit CPA pin CT 20 , generator sub-circuit PPG (FIG. 43) pin PPG 0 , sub-circuit AMP 1 pin GA 0 , sub-circuit AMP 2 pin GA 0 , capacitor C 4 , magnetic element BL 1 pin, transistor Q 6 source, transistor Q 9 source, sub-circuit SNA pin SNA 2 sub-circuit SNB pin SNB 2 , pin PF− FIG. 3 then to sub-circuit IPFFB pin PF−. Drain of output switch Q 1 is connected to diode D 4 anode, sub-circuit SN pin SNL 2 , then to pin 34 of FIG. 3A then to sub-circuit PFT 1 pin P 1 A. Snubber SN reduces the high voltage stress to Q 1 until flyback diode D 4 begins conduction. Additional rectification efficiency and protection is achieved by adding sub-circuit DSN (FIG. 30A) across flyback diode D 4 . Feedback corrected boost output voltage of the power factor corrected AC to DC converter stage appears across nodes PF+ and PF−. The regulated 385-volt boost output node PF+ connects to the following; sub-circuit SN pin SNOUT, diode D 4 cathode, sub-circuit IPFFB (FIG. 40) pin PF+, sub-circuit FBA pin PF+, then to pin PF+ of FIG. 3A, capacitors [C 16 ||C 17 ||R 17 ], magnetic element sub-circuit PTT 1 (FIG. 19) pin P 2 CT, snubber sub-circuit SNA (FIG. 31) pin SNA 3 , and snubber SNB (FIG. 32) pin SNB 3 , sub-circuit OVP pin PF+, capacitor C 4 and diode D 70 cathode. Magnetic element winding node S 1 H of sub-circuit PFT 1 is connected to pin 31 FIG. 3 then to sub-circuit CP pin CT 1 A and pin PFVC of sub-circuit PFB. Magnetic element winding node S 1 L of sub-circuit PFT 1 is connected to pin 33 FIG. 3 then to sub-circuit CP pin CT 2 A. Magnetic element winding node S 2 H of sub-circuit PFT 1 is connected to CPA pin CT 1 B. Magnetic element winding node S 2 L of sub-circuit PFT 1 is connected to CP pin CT 2 B. Sub-circuit PFB using feedback from the phase of the AC line, Q 1 switch current, magnetic bias first stage and output voltage feedback generates a command pulse on pin PFCLK. Pin PFCLK of sub-circuit PFB (FIG. 24) is connected to the input of buffer AMP amplifier pin GA 1 of sub-circuit AMP 1 . Buffered high-speed low impedance gate drive output pin GA 2 of sub-circuit AMP is connected to gate of switch FET Q 1 . The buffering provided by AMP shortens switch Q 1 “ON” and “OFF” times greatly reducing switch losses (See FIGS. 13 and 14 ). The source of Q 1 is connected to sub-circuit AMP pin GA 0 , pin 35 of FIG. 3A then to current sense resistor R 26 connected to return node BR−. The voltage developed across R 26 is fed back to PFB pin PFSC. This signal is used to protect the switch by reducing the pulse width in response to a low line or high load induced over current fault. The return line of sub-circuit FBA pin BR− is connected to node BR− and to pin BR− of sub-circuit PFB. This feedback is non-isolated; network values are selected for the first stage to develop a 385-Volt output at PF+. Sub-circuit feedback network FBA (FIG. 40A) pin PF 1 is connected to sub-circuit PFB pin PF 1 . Controller PFB modulates PFCLK signal to maintain a substantially constant 385-voltage at PF+ independent of line and load conditions. In the event of a component failure in sub-circuit FBA the PBF may command the converter to very high voltages. Sub-circuit OVP monitors the first stage boost in the event it exceeds 405-volts OVP will clamp the output of sub-circuit BR causing fuse F 1 in sub-circuit LLA to open. An alternate over voltage network OVP 1 (FIG. 42A) may replace OVP clamping the 18-volt control power stopping the boost action of the converter without opening the fuse. Sampled converter output at node from sub-circuit FBA pin PF 1 is connected to sub-circuit PFB pin PF 1 . The haversine on BR+ is used with an internal multiplier by PFB to generate variable width control pulses on pin PFCLK. The high frequency modulation of switch Q 1 makes the load/converter appear resistive to the AC line. Over temperature protection sub-circuit OTP pin TS+ is connected to sub-circuit AMP pin GA+. Thermal switch THS 1 is connected to Q 1 . In the event Q 1 reaches approximately 105C THS 1 opens removing power to sub-circuit AMP, safely shutting down the first stage. Normal operation resumes after the temperature decreases 20-30C closing THS 1 . The second stage is configured as a buck stage. It accepts the 385-Volt output of the first stage. By employing a second floating reference node PF− energy storage element capacitor C 4 the voltage to the final push-pull stage may be regulated with minimal loss. Power from sub-circuit CP pin CP 18 V+ is connected to pin 30 of FIG. 3A then to sub-circuit PWFM (FIG. 33) pin PWM+ and AMP 3 pin GA+. Feedback current from sub-circuit IPFFB pin FBC is connected to pin 36 FIG. 3A then to sub-circuit IFB pin FBC and sub-circuit PWFM pin PF 1 . Sub-circuit IPFFB only shunts current from this node if the output of the second stage is greater than 200-volts. When the converter reaches its designed output voltage, IFB shunts current from PWFM pin PF 1 signaling PWFM to reduce the pulse width on pin PWMCLK. Sub-circuit AMP 3 input pin is connected to sub-circuit PWFM pin PWMCLK. Output of AMP 3 buffer pin GA 2 is connected to gate of switch Q 2 . Drain of Q 2 is connected to anode of D 70 and non-saturating magnetic sub-circuit BL 1 pin P 2 B (FIG. 18 B). Turning on switch Q 2 charges C 4 also storing energy in magnetic element BL 1 . Releasing switch Q 2 allows energy stored in magnetic element BL 1 to charge C 4 through flyback diode D 70 . Larger pulse widths charge C 4 to larger voltages thus efficiently blocking part of the first stage voltage to the final push-pull stage. This action provides regulated voltage to the final converter stage. The third and final push-pull (transformer) converter stage provides the galvanic isolation, filtering and typically converts the internal high voltage bus to a lower regulated output voltage. The efficient push-pull stage produces alternating magnetizing currents in the NSME for maximum load over core mass. Constant frequency non-overlapping two-phase generator sub-circuit PPG (FIG. 43) generates the drive for the push-pull output stage. Phase one output pin PH 1 is connected to sub-circuit AMP 1 pin GA 1 , output pin PH 2 is connected to sub-circuit AMP 2 pin GA 1 . Output of amplifier buffer sub-circuit AMP 1 pin GAP 2 connects to gate of push-pull output switch Q 6 . Output of amplifier buffer sub-circuit AMP 2 pin GAP 2 connects to gate of push-pull output switch Q 9 . The buffering provided by AMP 1 and AMP 2 shortens switch Q 1 ON and OFF times greatly reducing switching losses. (See FIG. 13 and 14) Regulated 18-volt power from sub-circuit CPA pin CP 18 + is connected to amplifier buffer sub-circuit AMP 1 pin GA+, amplifier buffer sub-circuit AMP 2 pin GA+ and sub-circuit PPG pin PPG+. Drain of transistor Q 6 is connected to snubber network sub-circuit SNB pin SNB 1 and to non-saturating center tapped primary magnetic element sub-circuit PPT 1 pin P 2 H. Drain of transistor Q 9 is connected to snubber network sub-circuit SNA (FIG. 31) pin SNA 1 and sub-circuit PPT 1 pin P 2 L. Return node PF− connects source of transistor Q 6 to snubber network sub-circuit SNB pin SNB 3 , transistor Q 9 source, sub-circuit SNA pin SNA 3 and to return node GND 2 . Output of NSME sub-circuit PPT 1 pin SH connects to pin C 7 B of rectifier sub-circuit OUTA (FIG. 25 ), pin SL connects to C 8 B. center tap of PPT 1 pin SCT is the output return or negative node OUT− it connects to sub-circuit pin OUT− and sub-circuit IFB pin OUT− and RLOAD. Supply positive output from sub-circuit OUTA pin OUT+ is connected to RLOAD and sub-circuit IFB pin OUT+. Elements LL 1 , BR, PFA, AMP, Q 1 , IPFFB, IFB and PFT 1 provide power factor corrected AC to DC conversion and DC output regulation. The regulated high voltage output of this converter is used to power the efficient fixed frequency push-pull stages PPG, AMP 1 , AMP 2 , Q 6 , Q 9 , PPT 1 and OUTA. Magnetic element sub-circuit PPT 1 provides galvanic isolation and minimal voltage overshoot in the secondary thus minimizing filtering requirements of the rectifier sub-circuit OUTA. Sub-circuit IFB provides high-speed feedback to the AC DC converter, the speed of the boost stage provides precise output voltage regulation and active ripple rejection. In the event of a sudden line or load changes sub-circuit IPFFB compensates the internal boost. This system reduces losses by focusing output control in the middle (low current) stage of the converter and by using non-saturating magnetics, buffered switching, and rectifying snubbers throughout each stage. The combined improvements translate to higher system efficiencies, higher power densities, lower operating temperatures, and, improved thermal tolerance thereby reducing or eliminating the need for forced air-cooling per unit output. The non-saturating magnetic properties are relatively insensitive to temperature (see FIG. 17 ), thus allowing the converter to operate over a greater temperature range. In practice, the operating temperature for the Kool Mu NSME is limited to 200C by wire/core insulation; the non-saturating magnetic material remains operable to near its Curie temperature of 500C. This configuration provides power factor corrected input transient protection, rapid line-load and ripple compensation, excellent output regulation, output isolation and quiet efficient operation at high temperatures. FIG. 4 is a schematic diagram sub-circuit ACDCPF. FIG. 4 is a schematic diagram of a power factor corrected single stage AC to DC converter sub-circuit ACDCPF. The invention is comprised of line protection filter sub-circuit LL (FIG. 20) and full-wave rectifier sub-circuit BR (FIG. 22 ). A power factor corrected regulated boost stage with sub-circuits PFB (FIG. 24 ), snubber sub-circuit SN (FIG. 30 ), magnetic element sub-circuit PFT 1 A (FIG. 18 A), sub-circuit CP (FIG. 26 ), buffer sub-circuit AMP (FIG. 29 ), over temperature sub-circuit OTP (FIG. 28 ), and voltage feedback sub-circuit FBA (FIG. 40 A). Start up resistor R 2 , filter capacitor C 1 , PFC capacitor C 2 , flyback diode D 4 , switch transistor Q 1 , hold up capacitors C 17 and C 16 , and resistor R 17 . Table FIG. 4 Value/part Element number C1 .01 uf C2 1.8 uf R2 100k ohms R26 0.05 ohms Q1 IRFP 460 D4 STA1206 DI C17 100 uf C16 100 uf R17 375k ohms AC line is connected to sub-circuit LL (FIG. 20) between pins LL 1 and LL 2 . AC/earth ground is connected to node LL 0 . The filtered and voltage limited AC line appears on node/pin LL 5 of sub-circuit LL 1 and connected to node BR 1 of bridge rectifier sub-circuit BR (FIG. 22 ). The neutral/AC return leg of the filtered and voltage limited AC appears on pin LL 6 of sub-circuit LL is connected to input pin BR 2 of BR. The line voltage is full-wave rectified and is converted to a positive haversine appearing on node BR+ of sub-circuit BR (FIG. 22 ). Start up resistor R 2 connects BR+ to sub-circuit CP pin CP+. Node CP+ connects to pins PFA+ of power factor controller sub-circuit PFA (FIG. 24) and over temperature switch sub-circuit OTP (FIG. 28) pin GAP. Resistor R 2 provides start up power to the control element until the rectifier and regulator CP is at full output. Node S 1 H from PFT 1 A is connected to node PFVC sub-circuit PFB. The zero crossing of the core bias are sensed when the voltage at S 1 H is at zero. The core zero crossings are used to reset the PFC and start a new cycle. The positive node of the DC side of bridge BR+ is connected through capacitor C 2 to BR−. C 2 is selected for various line and load conditions to de-couple switching current from the line improving power factor. Primary of NSME sub-circuit PFT 1 A (FIG. 18A) pin P 1 B connects to pin SNL 1 of snubber sub-circuit SN (FIG. 30 ), sub-circuit PFB pin BR+ and connects to node BR+. The return line for the rectified AC power BR− is connected to the following pins; BR− of sub-circuit BR, sub-circuit PFB pin BR−, sub-circuit AMP pin GA 0 , sense resistor R 26 , capacitor [C 16 ||C 17 ||resistor R 17 ], capacitor C 2 , sub-circuit CP pin CT 0 , sub-circuit PFT 1 A pin S 1 CT and through EMI filter capacitor C 1 to earth ground node LL 0 . Drain of output switch Q 1 is connected to diode D 4 anode, sub-circuit PFT 1 A pin P 1 A and snubber sub-circuit SN pin SNL 2 . Additional rectification efficiency and protection is achieved by adding sub-circuit DSN (FIG. 30A) in parallel flyback diode D 4 . Sub-circuit provides reduces the high voltage stress to Q 1 until flyback diode D 4 begins conduction. Line coupled, power factor corrected boost regulated output voltage of the AC to DC converter stage (FIG. 1) appears on node PF+. The regulated boost output PF+ connects to the following; sub-circuit SN pin SNOUT, diode D 4 cathode, capacitor [C 16 ||C 17 ||R 17 ], and snubber DSN (FIG. 30A) pin SNOUT. Magnetic element winding node S 1 H of sub-circuit PFT 1 A is connected to CP pin CT 1 A and pin PFVC of sub-circuit PFB. Magnetic element winding node S 1 L of sub-circuit PFT 1 A is connected to CP pin CT 2 A. Sub-circuit PFB using the phase of the AC line, and load voltage generates a command pulse PFCLK. Pin PFCLK of sub-circuit PFB (FIG. 24) is connected to the input of buffer amplifier pin GA 1 of sub-circuit AMP 1 (FIG. 29 ). Buffered high-speed gate drive output pin GA 2 of sub-circuit AMP is connected to gate of switch FET Q 1 . The buffering provided by AMP shortens switch Q 1 ON and OFF times greatly reducing switch losses. The source of Q 1 is connected to current sense resistor R 26 , pin PFSC of sub-circuit PFB, connected then to return node BR−. The voltage developed across R 26 is feedback to PFB pin PFSC. This signal is used to protect the switch in the event of an over current fault. Thermal switch THS 1 is connected to Q 1 . In the event Q 1 reaches approximately 105C THS 1 opens removing power to sub-circuit AMP, safely shutting down the first stage. Normal operation resumes after the switch temperature drops 20-30C closing THS 1 . Sub-circuit feedback network FBA (FIG. 40A) pin PF 1 is connected to sub-circuit PFB pin PF 1 . Converter output at node PF+ (the junction of C 17 ||C 16 and D 4 ) is connected to sub-circuit FBA pin PF+. The return line of sub-circuit FBA pin BR− is connected to pin BR− of sub-circuit PFB. This feed back is non-isolated; network values are selected for a substantially constant 385-Volt output at PF+ relative to BR−. The high-voltage haversine from the rectifier section BR pin BR+ is connected to sub-circuit PFB pin BR+. The haversine is used with an internal multiplier by PFB to make the converter ACDCPF appear resistive to the AC line. Sub-circuits LL 1 , BR, PFB, AMP, Q 1 , OTP, FBA, IFB and PFT 1 A perform power factor corrected AC to DC conversion. The regulated high voltage output of this converter may be used use to power one or more external converters connected to the PF+ and BR− nodes. The NSME sub-circuit PPT 1 A provides efficient boost action at high power levels in a very small form factor. Sub-circuit FBA provides high-speed feedback to the converter the speed of the boost stage provides precise output voltage regulation and active ripple rejection. This configuration provides power factor corrected input transient protection, rapid line-load response, excellent regulation, and quiet efficient operation at high temperatures. FIG. 5 is a graph comparing typical currents in saturating and non-saturating magnetic elements. As the inductance does not radically change at high temperatures and currents in the NSME, the large current spikes due to the rapid reduction of inductance common in saturating magnetics is not seen. As a result, destructive current levels, excessive gap leakage, magnetizing losses, and magnetic element heating are avoided in NSME. FIG. 6 is a schematic for non-isolated low side switch buck converter sub-circuit NILBK. Sub-circuit NILBK consists of resistor R 20 , diode D 6 , capacitor C 6 , FET transistor Q 111 , sub-circuit CP (FIG. 26 ), sub-circuit PFT 1 A (FIG. 18 A), sub-circuit IFB (FIG. 40 B), sub-circuit AMP (FIG. 29) and sub-circuit PWFM (FIG. 33 ). FIG. 6 Table Element Value/part number R20 100k ohms R20 STA1206 DI Q111 IRFP460 C6 10 uf External power source VBAT connects to pins DCIN+ and DCIN−. From DCIN+ through resistor R 20 connects to sub-circuit CP pin CP+, sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Resistor R 20 provides startup power to the converter before regulator sub-circuit CP reaches it full 18-volt output. VBAT negative is connected to pin DCIN− connects to sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , Q 111 source, sub-circuit IFB pin FBE, sub-circuit CP pin CT 0 , and sub-circuit PFT 1 pin S 1 CT. Magnetic element winding node S 1 H of sub-circuit PFT 1 A is connected to CP pin CT 1 A. Magnetic element winding node S 1 CT of sub-circuit PFT 1 is connected to CP pin CT 0 . Magnetic element winding node S 1 H of sub-circuit PFT 1 A is connected to CP pin CT 2 A. The regulated 18 volts from sub-circuit CP+ is connected to R 20 , sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit PWFM is designed for variable pulse width operation. PWFM is configured for maximum pulse width 90-95% with no feedback current from sub-circuit IFB pin FBC. Increasing the feedback current reduces the pulse-width and output voltage from converter NILBK. Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP. The output of sub-circuit AMP pin GA 2 is connected to the gate of Q 111 . Input node DCIN+ connects to the cathode of flyback diode D 6 , sub-circuit IFB pin OUT+, resistor RLOAD, capacitor C 6 and pin B+. The drain of Q 111 is connected to sub-circuit PFT 1 pin P 1 B and the anode of D 6 . Pin P 1 A of sub-circuit PFT 1 A is connected to capacitor C 6 , RLOAD, sub-circuit IFB pin OUT− and to node B−. With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 111 . Switch Q 111 conducts charging capacitor C 10 through NSME PFT 1 A from source VBAT and storing energy in PFT 1 A. Feedback output pin FBC from sub-circuit IFB is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . Sub-circuit IFB removes current from PW 1 commanding PWFM to reduce the pulse-width or on time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PWFM switches output pin CLK low turning “off” Q 111 stopping the current into PFT 1 A. The energy not transferred into regulator sub-circuit CP load is released from NSME PFT 1 A into the now forward biased diode D 6 charging capacitor C 6 . By modulating the “ont” time of switch Q 111 the converter buck voltage is regulated. Regulated voltage is developed across Nodes B− and B+. Sub-circuit IFB provides the isolated feedback voltage to the sub-circuit PWFM. When sub-circuit IFB senses the converter output (nodes B+ and B−) is at the designed voltage, current from REF is removed from PM 1 . Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 111 is removed stopping all buck activity capacitor C 6 discharges through RLOAD. Input current from VBAT is sinusoidal making the converter very quiet. In addition the switch Q 111 is not exposed to large flyback voltage. Placing less. stress on the switches thereby increasing the MTBF. Sub-circuit NILBK takes advantage of the desirable properties of the NSME in this converter topology. Adjusting the NSME 100 (FIG. 18A) primary inductance and component values in sub-circuit IFB determines the output buck voltage. FIG. 8 is a schematic for a tank coupled single stage converter sub-circuit TCTP. Sub-circuit TCTP consists of resistor R 20 and RLOAD, capacitor C 10 , Darlington transistors Q 10 and Q 20 , sub-circuit CP (FIG. 26 ), sub-circuit PFT 1 (FIG. 18 ), sub-circuit OUTB (FIG. 25 A), sub-circuit IFB (FIG. 40B) and sub-circuit PWFM (FIG. 33 ). FIG. 8 Table Element Value/part number R20 5k ohms Q10 TST541 Q20 IRFP460 C10 1.8 uf External power source VBAT connects to pins DCIN+ and DCIN−. From DCIN+ connects to Q 10 collector then through resistor R 20 connects to sub-circuit CP pin CP+ and to sub-circuit PWFM pin PWFM+. Resistor R 20 provides startup power to the converter before regulator sub-circuit CP reaches it full 18-volt output. VBAT negative is connected to pin DCIN− ground/return node GND. Node GND connects to sub-circuit PWFM 0 pin PWFM 0 , Q 20 collector, C 10 , sub-circuit CP pin CT 0 and sub-circuit PFT 1 pin S 1 CT. Magnetic element winding node S 1 H of sub-circuit PFT 1 is connected to CP CT 1 A. Magnetic element winding node S 1 L of sub-circuit PFT 1 is connected to CP CT 2 A. Magnetic element winding node S 1 CT of sub-circuit PFT 1 is connected to CP pin CT 0 . Magnetic element winding node S 2 H of sub-circuit PFT 1 is connected to CP pin CT 2 A. The regulated 18 volts from sub-circuit CP+ is connected to R 20 and to sub-circuit PWFM pin PWFM+. Sub-circuit PWFM is designed for a constant 50% duty cycle variable frequency generator. Sub-circuit PWFM clock output pin CLK is connected to the base of Q 10 and Q 20 . The emitters of Q 10 and Q 20 are connected to sub-circuit PFT 1 pin P 1 B. This forms an emitter follower configuration. Pin P 1 A of sub-circuit PFT 1 is connected through tank capacitor C 10 to node GND. With PWFM CLK pin high forward biased transistor Q 10 supplies current to the tank from BAT 1 charging capacitor C 10 through NSME PFT 1 and transferring energy into PFT 1 . Sub-circuit PWFM switches CLK low turning “off” Q 10 stopping the current into PFT 1 . Energy not transferred into the load is released from NSME PFT 1 into the now forward biased PNP transistor Q 20 back into capacitor C 10 . Thus any energy not used by the secondary loads is transferred back to the primary tank to be used next cycle. When the switching occurs at the resonant frequency large circulating currents develop in the tank. Also C 10 is charged and discharged to very large voltages. Oscillograph in FIG. 35 is the actual voltage developed across capacitor C 10 with VBAT equal to 18 volts. A very large 229-Volts peak to peak was developed across the nodes P 1 A and P 1 A of NSME PFT 1 . The large primary voltage generates large biases in the NSME PFT 1 to be flux harvested by the windings 102 and 103 (FIG. 18) and transferred to a load or rectifier sub-circuit OUTB. Magnetic element winding node S 2 L of sub-circuit PFT 1 is connected to OUTB C 8 b . Magnetic element winding node S 2 H of sub-circuit PFT 1 is connected to C 7 B of sub-circuit OUTB node OUT−. Node OUT− is connected to RLOAD, pin B− and to sub-circuit IFB pin OUT−. Rectified power is delivered to pin OUT+ of OUTB and is connected to RLOAD, pin B+ and to sub-circuit IFB pin OUT+. Sub-circuit IFB provides the isolated feedback signal to the sub-circuit PWFM. Frequency control pin FM 1 of sub-circuit PWFM is connected to sub-circuit IFB pin FBE. Internal reference pin REF of sub-circuit PWFM is connected to sub-circuit IFB pin FBC. PWFM is designed to operate at the resonate frequency of the tank. When sub-circuit IFB senses the converter output is at the designed voltage, current from REF is injected into FM 1 . Injecting current into FM 1 commands PWFM to a lower frequency. Operating below resonance reduces the amount of energy added to the primary tank thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM to 0 Hz all primary activity stops. Input current from VBAT is sinusoidal making the converter very quiet. In addition the switches Q 10 and Q 20 are never exposed to the large circulating voltage (FIG. 35 ). Placing less stress on the switches thereby increasing the MTBF. Sub-circuit TCTP takes advantage of the desirable properties of the NSME in this converter topology. Adjusting secondary turns allows TCTP to generate very large AC or DC output voltages as well as low-voltage high current outputs. FIG. 9 is a schematic for non-isolated low side switch boost converter sub-circuit NILSBST. Sub-circuit NILSBST consists of resistor R 20 and RLOAD, diode D 6 , capacitor C 6 , FET transistor Q 111 , sub-circuit CP (FIG. 26 ), sub-circuit PFT 1 A (FIG. 18 A), sub-circuit FBI (FIG. 41 ), sub-circuit AMP (FIG. 29) and sub-circuit PWFM (FIG. 33 ). FIG. 9 Table Element Value/part number R20 100 k ohms Q111 IRFP460 D6 STA1206 DI C6 200 uf External power source VBAT connects to pins DCIN+ and DCIN−. From DCIN+ Resistor R 20 connects to sub-circuit CP pin CP+, sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Resistor R 20 provides startup power to the converter before regulator sub-circuit CP reaches it full 18-volt output. VBAT negative is connected to pin DCIN− and ground return node GND. Node GND connects to sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , Q 111 source, sub-circuit FBA pin BR−, sub-circuit FBA pin FBA, sub-circuit CP pin CT 0 , capacitor C 6 , resistor RLOAD, transistor Q 111 source, and sub-circuit PFT 1 pin S 1 CT. Magnetic element winding node S 1 H of sub-circuit PFT 1 A is connected to CP pin CT 1 A. Magnetic element winding node S 1 CT of sub-circuit PFT 1 is connected to CP pin CT 0 . Magnetic element winding node S 2 H of sub-circuit PFT 1 A is connected to CP pin CT 2 A. The regulated 18 volts from sub-circuit CP+ is connected to R 20 , sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit PWFM is designed for variable pulse width operation. PWFM is configured for maximum pulse width 90-95% (maximum boost voltage) with no feedback current from sub-circuit FBI. Increasing the feedback current reduces the pulse-width reducing the boost voltage and reducing the output from converter NILSBST. Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP. The output of sub-circuit AMP pin GA 2 is connected to the gate of Q 111 . Input node DCIN+ connects to the NSME PFT 1 A pin P 1 A. The drain of Q 11 is connected to sub-circuit PFT 1 A pin P 1 B and the anode of D 6 . Cathode of diode D 6 is connected to sub-circuit FBA pin PF+, resistor RLOAD, C 6 and pin BK+. With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 111 . Switch Q 111 conducts reverse biasing diode D 6 capacitor C 10 stops charging through NSME PFT 1 A from source VBAT. During the time Q 111 is conducting, energy is stored in NSME sub-circuit PFT 1 A. Feedback output pin FBC from sub-circuit FBI is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . Sub-circuit FBI removes current from PW 1 commanding PWFM to reduce the pulse-width or on time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PFFM switches CLK low turning “off” Q 111 stopping the current into PFT 1 A. The energy not transferred into regulator sub-circuit CP load is released from NSME PFT 1 A into the now forward biased diode D 6 charging capacitor C 6 . By modulating the “on” time of switch Q 111 the converter boost voltage is regulated. Regulated voltage is developed across Nodes B− and B+. Sub-circuit IFB provides the feedback current to the sub-circuit PWFM. When sub-circuit IFB senses the converter output (nodes B+ and B−) is at or greater than the designed voltage, current is removed from PM 1 . Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 111 is removed stopping all boost activity capacitor C 6 charges to VBAT. Input current from VBAT is sinusoidal making the converter very quiet. In addition the switch Q 111 is not exposed to large flyback voltage. Placing less stress on the switches thereby increasing the MTBF. Sub-circuit NILBK takes advantage of the desirable properties of the NSME in this converter topology. Adjusting the NSME 100 (FIG. 18A) primary inductance and component values in sub-circuit IFB determines the output boost voltage. FIG. 10 is a schematic for a two stage isolated DC to DC boost controlled push-pull converter BSTPP. Sub-circuit BSTPP consists of diode D 14 , capacitor C 14 , FET transistor Q 14 , sub-circuit REG (FIG. 36 ), sub-circuit BL 1 (FIG. 18 B), sub-circuit IFB (FIG. 40 B), sub-circuit AMP (FIG. 29 ), sub-circuit DCAC 1 , and sub-circuit PWFM (FIG. 33 ). External power source VBAT connects to pins DCIN+ and DCIN−. FIG. 10 Table Element Value/part number Q31 IRFP460 D14 STA1206 DI C14 10 uf From pin DCIN+ connects to sub-circuit REG pin RIN+ and sub-circuit BL 1 pin P 1 A. Voltage regulator sub-circuit output pin + 18 V connects to sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit REG provides regulated low voltage power to the controller and to the main switch buffers. VBAT negative is connected to pin DCIN− and ground return node GND. Node GND connects to sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , Q 14 source, capacitor C 14 , sub-circuit IFB pin FBE, sub-circuit REG pin REG 0 , sub-circuit DCAC 1 pin DC−. Sub-circuit PWFM (FIG. 33) is designed for variable pulse width operation. The nominal frequency is between 20-600 Khz PWFM is configured for maximum pulse width 90% (maximum boost voltage) with no feedback current from sub-circuit FBI. Increasing the feedback current reduces the pulse-width reducing the boost voltage and reducing the output from converter BSTPP. Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP (FIG. 29 ). The output of switch speed up buffer sub-circuit AMP pin GA 2 is connected to the gate of Q 14 . Input node DCIN+ connects to the NSME BL 1 pin P 1 A. The drain of Q 14 is connected to sub-circuit BL 1 pin P 1 B and the anode of D 14 . Cathode of flyback diode D 14 is connected to sub-circuit DCAC 1 pin DC+ and C 14 . With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 14 . Switch Q 14 conducts reverse biasing diode D 14 capacitor C 14 stops charging through NSME BL 1 from source VBAT. During the time Q 14 is conducting, energy is stored in NSME sub-circuit BL 1 . Feedback output pin FBC from sub-circuit IFB is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . Sub-circuit IFB removes current from PW 1 commanding PWFM to reduce the pulse-width or “on” time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PFFM switches CLK low turning “off” Q 14 stopping the current into BL 1 . The energy is released from NSME BL 1 into the now forward biased flyback diode D 14 charging capacitor C 14 . By modulating the “on” time of switch Q 14 the converter boost voltage is regulated. Regulated voltage is developed across C 14 Nodes DC+ and GND is provided to the isolated constant frequency push-pull DC to AC converter sub-circuit DCAC 1 (FIG. 2 ). Sub-circuit DCAC 1 provides efficient conversion of the regulated boost voltage to a higher or lower voltage set by the magnetic element-winding ratio. The center tap of the push-pull output magnetic is connected to, sub-circuit OUTB pin OUT−, RLOAD, sub-circuit IFB pin OUT− and the pin OUT− forming the return line for the load and feedback network. Output of sub-circuit DCAC 1 in ACH is connected to sub-circuit OUTB pin C 7 b . Output of sub-circuit DCAC 1 pin ACL is connected to sub-circuit OUTB pin C 8 b . Sub-circuit OUTB provides rectification of the AC power generated by sub-circuit DCAC 1 . Since the non-saturating magnetic converter has low output ripple, minimal filtering is required by OUTB. This further reduces cost and improves efficiency as losses to filter components are minimized. Sub-circuit IFB provides the isolated feedback current to the sub-circuit PWFM. When sub-circuit IFB senses the converter output (nodes OUT+ and OUT−) is greater than the designed/desired voltage, current is removed from node PM 1 . Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 14 is removed stopping all boost activity capacitor C 14 charges to VBAT. As the non-saturating does not saturate the destructive noisy current “spikes” common to prior art are absent. Input current from VBAT to charge C 14 is sinusoidal making the converter very quiet. In addition the switch Q 14 is not exposed a potentially destructive current spike. Placing less stress on the switches thereby increasing the MTBF. Sub-circuit BSTPP takes advantage of the desirable properties of the NSME. Adjusting the NSME BL 1 (FIG. 18B) sets the amount of boost voltage available to the final push-pull isolation stage. Greater efficiencies are achieved at higher voltages. The final output voltage is set by the feedback set point and the turns ratio of the push-pull element PPT 1 (FIG. 19 ). FIG. 11 is a graph of permeability as a function of temperature for typical prior art magnetic element material. The high permeability material in FIG. 11 exhibits large changes in permeability of almost 100% over a 100C range as compared to the less than 5% change for the material in FIG. 17 . The increase in permeability at high temperatures of the prior art material increases the flux density resulting in core saturation for a constant power level. (See FIG. 12) Thus the prior art core must be derated at least 100% to operate over extended temperatures. The instant invention takes advantage of the desirable properties of the NSME. Eliminating the need to derate the magnetic element. As the magnetic element performs better at high temperatures, currently limited by melting wire insulation. FIG. 12 is a graph of flux density as a function of temperature for typical prior art magnetic element material. The reduction of maximum flux density with temperature is typical of the saturating magnetic element prior art material. Thus the prior art core is commonly derated at least 100% to operate over extended temperatures. Resulting in a larger more expensive design, and or the requirement to cool the core. FIG. 12A is a graph of magnetic element losses for various flux densities and operating frequencies typical of prior art magnetic element material. FIG. 13 is a graph showing standard switching losses. The hashed area represents the time when the switch is in a resistive state. The hashed area is proportional to the amount of energy lost each time the output switch operates. Total power lost is the product of the loss per switch times the switching frequency. FIG. 14 is a graph showing the inventions switching losses. The hashed area represents the time when the switch is in a resistive state. The smaller hashed area is due to the action of the buffer in FIG. 29 and the snubber isolation diode D 805 in FIG. 30 . Generally the NSME has a wider usable frequency band and can be magnetized from higher primary voltages. Higher operating voltages have proportionally smaller currents for a given power level thus proportionally lower losses. Switching losses more closely resemble I 2 R losses. Most switching loss occurs during turn “on” and turn “off” transitions; total switching losses are reduced proportionally by the lower switching frequencies and faster transition times characteristic of the disclosed NSME converters. In addition the properties of the NSME allow operation at temperature extremes beyond the tolerance of standard prior art magnetics and their geometry's. The combined contributions of the above yields a converter that requires little or no forced air-cooling. (See FIG. 15, 16 , and 17 ) FIG. 15 is a graph of the NSME magnetization curves for Kool Mu material. The invention makes advantageous use of the available saturation range of the NSME. FIG. 16 is a graph of the Kool Mu NSME losses for various flux densities and operating frequencies. It can be seen from the data that much higher flux densities are available per unit losses over the prior art. FIG. 17 is a plot of permeability vs. temperature for several Kool Mu materials. This data demonstrates the usefulness and stability of the magnetic properties over temperature. FIG. 18 is a schematic representation of the non-saturating magnetic boost element PFT 1 . Sub-circuit PFT 1 consists of a primary winding 100 around a NSME 101 with two center-tapped windings 102 and 103 . FIG. 18 Table Element Value/part number 100 55 turns 203 uh 101 2 × 77932-A7 102 14 turns 102 14 turns The primary winding 100 has nodes P 1 B and P 1 A for connections to external AC source. Secondary 102 winding has center-tapped node S 1 CT and node S 1 H and S 1 L connections to the upper and lower halves respectively. Secondary 103 winding has center-tapped node S 2 CT and node S 2 H and S 2 L connections to the upper and lower halves respectively. Both 102 and 103 are connected to external full-wave rectifier assemblies. Magnetic element magnetic element 101 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which ranges from 1500 to 5000 u. Flyback management is of concern when using NSME in a boost converter because the magnetic element generates high drain source voltages across the primary switch during the reverse recovery time of the flyback (output) diode. The magnitude per cycle of flyback current from NSME is greater for a given input magnetizing force relative to the prior art. (See FIG. 5) For example, Kool Mu torroids (Materials from Magnetics) are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight, 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air, (permeablity=1); a molypermalloy powder, (MPP) a high flux MPP, a powder, a gapped ferrite, a tape wound, a cut magnetic element, a laminated, or an amorphous magnetic element. Unlike the prior art, the NSME is temperature tolerant in that the critical parameters of permeability and saturability remain substantially unaffected during extreme thermal operation over time. Some materials such as air also exhibit little or no change in permeability or saturation levels over time, temperature, and conditions. The prior art uses high permeability saturable materials often greater than 2000 u permeability. These magnetics exhibit undesirable changes in permeability and saturation during operation at or near rated output making operation at high power levels and temperature difficult. See the permeability vs. temperature FIG. 11 . This deficiency is overcome by the use of expensive oversized magnetic elements or output current sharing with multiple supplies. (See the graph b sat VS. temperature FIG. 12) This invention takes advantage of the desirable properties of NSME. See the permeability vs. temperature FIG. 17 . Prior art saturating magnetic element commonly is operating at frequencies greater than 500 KHz to achieve greater power levels. As a result practitioners experience exponentially greater core losses (see FIG. 12A) at high frequencies. NSME support operation at lower frequencies 20-600 KHz further reducing switching losses and magnetic element losses allowing operation at even higher temperatures. See the loss density vs. flux density FIG. 16 . Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I2R/copper losses. The magnetic element is operated at duty cycles of 0%+ to 90%, which, when used to control the primary side push-pull voltage, results in efficiencies on the order of 90%. FIG. 18A is a schematic representation of the NSME PFT 1 A Sub-circuit transformer PFT 1 A consists of a primary winding 100 around a NSME 101 with a center-tapped winding 102 . FIG. 18A Table Element Value/part number 100 55 turns 230 uh 101 2 × 77932-A7 102 14 turns The primary winding 100 has nodes P 1 B and P 1 A for connections to external AC source. Secondary 102 winding has center-tapped node S 1 CT and node S 1 H and S 1 L connections to the upper and lower halves respectively. Winding 102 are typically connected to external full-wave rectifier assemblies. Magnetic element 101 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which is on the order of 2500 u. Flyback management is of concern when using such a magnetic element because the magnetic element generates high drain source voltages across the primary switch during the reverse recovery time of the flyback diode. Flyback current is available for longer periods after the primary switch opens. (See FIG. 5) For example, Kool Mu (Materials from Magnetics are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight: 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air; (air magnetic element permeablity=1); a molypermalloy powder (MPP) magnetic element; a high flux MPP magnetic element; a powder magnetic element; a gapped ferrite magnetic element; a tape wound magnetic element; a cut magnetic element; a laminated magnetic element; or an amorphous magnetic element. During operation the temperature of the NSME rises, the permeability slowly decreases, thereby increasing the saturation point. Some materials such as air exhibit no or very small changes in permeability or saturation levels. Unlike prior art using high permeability materials greater than 2000 u permeability rapidly increases at high temperatures. See the permeability vs. temperature FIG. 11 . Prior art also suffers from reduced magnetic element saturation levels at high temperatures, making operation at high power levels and temperature difficult and may require the use of expensive oversized magnetic elements. See the graph b sat vs. temperature FIG. 12 this invention takes advantage of the desirable NSME properties. See the permeability vs. temperature FIG. 17 . Operation at lower frequencies 20-600 KHz reduces switching losses and magnetic element losses allowing operation at higher temperatures. See the loss density vs. flux density FIG. 16 . Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I2R/copper losses. The magnetic element is operated at duty cycles of 0%+ to 90%, which, when used to control the primary side push-pull voltage, results in efficiencies on the order of 90%. FIG. 18B is a schematic representation of the NSME BL 1 Sub-circuit BL 1 consists of a winding 100 around a NSME 101 . FIG. 18 Table Element Value/part number 107 40 turns 50 uh 101 2 × 77932-A7 Magnetic element BL 1 may also be constructed from one or more magnetic elements in series or parallel. Assuming minimal mutual coupling the total inductance is the arithmetic sum of the individual inductances. For elements in parallel the (assuming minimal mutual coupling) the total inductance is the reciprocal of the arithmetical sum of the reciprocal of the individual inductances. In this way multiple magnetic elements may be arranged to meet packaging, manufacturing, and power requirements. The primary winding 100 has nodes P 2 B and P 2 A for connections to external AC source. Magnetic element 101 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which is on the order of 2500 to 5000 u. Flyback management is of concern when using such a magnetic element because the magnetic element generates high drain source voltages across the primary switch during the reverse recovery time of the flyback diode. Flyback current is available for longer periods after the primary switch opens. (See FIG. 5) For example, Kool Mu (Materials from Magnetics are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight: 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air (air magnetic element permeablity=1); a molypermalloy powder (MPP) magnetic element; a high flux MPP magnetic element; a powder magnetic element; a gapped ferrite magnetic element; a tape wound magnetic element; a cut magnetic element; a laminated magnetic element; or an amorphous magnetic element. During operation temperature of the magnetic element rises, the permeability slowly decreases, thereby increasing the saturation point. Some materials such as air exhibit no or very small changes in permeability or saturation levels. Unlike prior art using high permeability materials greater than 2000 u permeability rapidly increases at high temperatures. See the permeability vs. temperature FIG. 11 . Prior art also suffers from reduced magnetic element saturation levels at high temperatures, making operation at high power levels and temperature difficult and may require the use of expensive oversized magnetic elements. (See the graph b sat vs. temperature FIG. 12) This invention takes advantage of the desirable NSME properties. (See the permeability vs. temperature FIG. 17.) Prior art often operates at high switching frequencies 100-1000 kHz to avoid the saturation problem. Only to increase switching and core losses. (See FIG. 12A) This inventions use of the desirable NSME properties allows operation at lower frequencies 20-600 KHz further reducing switching losses and magnetic element. See the loss density vs. flux density FIG. 16 . Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I2R/copper losses. FIG. 18C is a schematic representation of a distributed NSME PFT 1 D. This is shown to exemplify distributed magnetics enable advantageous high voltage converter design variations that support form factor flexibility and multiple parallel secondary outputs from series coupled voltage divided primary windings across multiple NSME. This magnetic strategy is useful in addressing wire insulation, form factor and packaging limitations, circuit complexity and manufacturability. In this example a 500W converter is required to fit in a low profile package. Sub-circuit PFTD 1 consists of three magnetic elements 120 , 121 and 124 with series connected primaries. FIG. 18C Table Element Value/part number 113 77352-A7 122 23 Turns 123 23 Turns 112 67 uh (55 turns) 114 77352-A7 116 67 uh (55 turns) 117 77352-A7 118 67 uh (55 turns) AC voltage is applied to 112 pin P 1 B then from P 1 C through conductor 115 to 116 pin P 1 D. Winding 116 pin P 1 E is connected through conductor 119 to 118 pins P 1 F then to pin P 1 A. Original Sub-circuit PFT 1 (FIG. 18) consists of a primary winding 100 around a NSME 101 with two center-tapped windings 122 and 123 . By way of example sub-circuit PFT 1 D will be implemented as three magnetic elements. For a 500-watt expression a total inductance of 203 uH is required in winding 100 (FIG. 18 ). Dividing the primary inductance by the number of elements, in this case three requires elements 112 , 116 and 118 have 67 uH of inductance. The energy storage is equally distributed over the magnetic assembly 120 , 121 and 124 . The 500 watt coverter in (FIG. 1) employs two (Kool Mu part number 77932-A7) 0.9 oz (25 gram) NSME to form 101 (FIG. 18 ). Sub-circuit PFT 1 magnetic element 101 (FIG. 18) may be expressed as three 0.5-0.7 oz (14-19 gram) elements. Three 0.5-oz Kool Mu elements (part number 77352-A7) were selected. To realize 67 uH of primary inductance 55 turns are required for elements 112 , 116 and 118 . The primary circuit has nodes P 1 B and P 1 A for connections to external AC source. Secondary 102 winding has center-tapped node S 1 CT and node S 1 H and S 1 L connections to the upper and lower halves respectively. Secondary 123 winding has center-tapped node S 2 CT and node S 2 H and S 2 L connections to the upper and lower halves respectively. Both 122 and 123 are connected to external full-wave rectifier assemblies. Magnetic element magnetic element 120 , 121 and 124 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which is on the order of 2500 u. Flyback management is of concern when using such a magnetic element because the magnetic element generates high drain source voltages across the primary switch during the reverse recovery time of the flyback diode. Flyback current is available for longer periods after the primary switch opens. (See FIG. 5) For example, Kool Mu (Materials from Magnetics are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight: 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air (air magnetic element permeablity=1); a molypermalloy powder (MPP) magnetic element; a high flux MPP magnetic element; a powder magnetic element; a gapped ferrite magnetic element; a tape wound magnetic element; a cut magnetic element; a laminated magnetic element; or an amorphous magnetic element. During operation the temperature of the NSME, the permeability slowly decreases, thereby increasing the saturation point. Some materials such as air exhibit no or very small changes in permeability or saturation levels. Unlike prior art using high permeability materials greater than 2000 u permeability rapidly increases at high temperatures. See the permeability vs. temperature FIG. 11 . Prior art also suffers from reduced magnetic element saturation levels at high temperatures, making operation at high power levels and temperature difficult and may require the use of expensive oversized magnetic elements. (See the graph b sat vs. temperature FIG. 12) This invention takes advantage of the desirable NSME properties. See the permeability vs. temperature FIG. 17 . Prior art saturating magnetic element commonly is operating at frequencies greater than 500 KHz to achieve greater power levels. As a result practitioners experience exponentially greater core losses (see FIG. 12A) at high frequencies. NSME allows operation at lower frequencies 20-600 KHz further reduces switching losses and magnetic element losses allowing operation at even higher temperatures. (See the loss density vs. flux density FIG. 16) Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I 2 R/copper losses. The magnetic element is operated at duty cycles of 0%+ to 90%, which, when used to control the primary side push-pull voltage, results in efficiencies on the order of 90%. FIG. 19 is a schematic representation of the non-saturating push-pull magnetic element sub-circuit PPT 1 Sub-circuit PPT 1 consists of a center-tapped primary winding 104 around a NSME 106 with one secondary center-tapped winding 105 . FIG. 19 Table Element Value/part number 106 77259-A7 105 10 Turns 104 70 Turns The primary winding 104 has nodes P 2 H and P 2 L for connections to external AC sources, and common center-tap node P 2 CT. Secondary 105 winding has center-tapped node SCT and nodes SH and SL connections to the upper and lower halves respectively. The invention is not limited to a single output. More secondary windings may be added for additional outputs. Secondary 105 is connected to an external full-wave rectifier assembly (Example FIG. 25 or 26 ). The magnetic element magnetic element 106 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which is on the order of 2500 u. Flyback management is of concern when using such a magnetic element as high drain source voltages across the primary switch are generated during the reverse recovery of the flyback diode. The falling flyback current is available for a longer period. (See FIG. 5) For example, Kool Mu (magnetic elements from Magnetics are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight; 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air (comprise an air magnetic element); a molypermalloy powder (MPP) magnetic element; a high flux MPP magnetic element; a powder magnetic element; a gapped ferrite magnetic element; a tape wound magnetic element; a cut magnetic element; a laminated magnetic element; or an amorphous magnetic element. During operation the temperature of the NSME rises, the permeability slowly decreases, thereby increasing the saturation point. Unlike prior art using high permeability materials greater than 2000 u permeability rapidly increases at high temperatures. See the permeability vs. temperature FIG. 11 . Prior art also suffers from reduced magnetic element saturation levels at high temperatures, making operation at high power levels and temperature difficult and may require the use of expensive oversized magnetic elements. (See the b sat vs. temperature FIG. 12) This invention takes advantage of the desirable NSME properties. (See the permeability vs. temperature FIG. 17) Operation at lower frequencies 20-600 KHz reduces switching losses and magnetic element losses allowing operation at higher temperatures. See the loss density vs. flux density FIG. 16 . Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I2R/copper losses. The magnetic element primary is driven in a push-pull fashion at a duty cycle of 48-49% resulting in efficient use of the magnetic element volume. FIG. 19A is a schematic representation of the non-saturating push-pull magnetic element sub-circuit PPT 1 . Sub-circuit PPT 1 consists of a center-tapped primary winding 134 around a NSME 136 with one secondary center-tapped winding 135 . FIG. 19A Table Element Value/part number 136 77259-A7 135 10 Turns 134 70 Turns The primary winding 134 has nodes P 2 H and P 2 L for connections to external AC sources, and common center-tap node P 2 CT. Secondary 135 winding has center-tapped node SCT and nodes SH and SL connections to the upper and lower halves respectively. The invention is not limited to a single output winding. More secondary windings may be added for additional outputs. Secondary 135 is connected to an external full-wave rectifier assembly such as OUTA (FIG. 25 ), OUTB (FIG. 25A) and OUTBB (FIG. 25 B). The magnetic element 136 comprises a non-saturating, low permeability magnetic material. The permeability is on the order of 26 u with a range of 1 u to 550 u, as compared to the prior art, which is on the order of 2500 u. Flyback management is of concern when using such a magnetic element as high drain source voltages across the primary switch are generated during the reverse recovery of the flyback diode. The falling flyback current is available for a longer period. (See FIG. 5) For example, Kool Mu (magnetic elements from Magnetics are suitable for this application. This material is not identified by way of limitation. The material comprises, by weight; 85% iron, 6% aluminum, and 9% silicon. Further, the magnetic element may be air (comprise an air magnetic element); a molypermalloy powder (MPP) magnetic element; a high flux MPP magnetic element; a powder magnetic element; a gapped ferrite magnetic element; a tape wound magnetic element; a cut magnetic element; a laminated magnetic element; or an amorphous magnetic element. During operation the temperature of the NSME rises, the permeability slowly decreases, thereby increasing the saturation point. Unlike prior art using high permeability materials greater than 2000 u permeability rapidly increases at high temperatures. See the permeability vs. temperature FIG. 11 . Prior art also suffers from reduced magnetic element saturation levels at high temperatures, making operation at high power levels and temperature difficult and may require the use of expensive oversized magnetic elements. (See the bsat vs. temperature FIG. 12) This invention takes advantage of the desirable NSME properties. (See the permeability vs. temperature FIG. 17) Operation at lower frequencies 20-600 KHz reduces switching losses and magnetic element losses allowing operation at higher temperatures. See the loss density vs. flux density FIG. 16 . Unlike the prior art, the instant invention uses voltage mode control with over-current shutdown. Material selection is also based upon mass and efficiency. By increasing the mass of the magnetic element, more energy is coupled more efficiently. Since there are reduced losses, the dissipation profile follows I2R/copper losses. The magnetic element primary is driven in a push-pull fashion at a duty cycle of 48-49% resulting in efficient use of the magnetic element volume. FIG. 20 Lightning Input Protection and Filter Sub-circuit LL FIG. 20 is a schematic showing the inventions filter and lightning input protection circuit for an AC line connected converter. The protection sub-circuit LL comprises a Spark gap A 1 , diodes D 20 and D 21 , capacitor C 1 and magnetic elements L 1 and L 2 . FIG. 20 Table Element Value/part number L1 375 uH L2 375 uH C61 0.01 uF C60 0.01 uF A1 400 V Spark Gap C1 0.1 uF D20 1000 V/25A D21 1000 V/25A D22 1000 V/25A D23 1000 V/25A C2 1.8 uf The AC line is connected to node LL 2 . The common input frequencies of DC to 440 Hz may be extended beyond this range with component selection. Node LL 2 is connected to NSME L 1 then to node LL 5 , the spark gap A 1 , anode of diode D 22 and the cathode of diode D 20 . Filter capacitor C 60 is connected between node LL 0 and LL 6 . Filter capacitor C 61 is connected between node LL 0 and LL 5 . The low side of AC line is connected to node LL 1 then to magnetic element L 2 the other side L 2 is connected to spark gap A 1 , anode of diode D 23 and the cathode of diode D 21 and to node LL 6 . Capacitor C 1 is connected to earth ground C 1 attenuates noise generated by the converter. The use of non-saturating magnetic allows the input magnetic elements to absorb very large voltages and currents commonly generated by lightning, often without causing spark gap A 1 to clamp. During UL testing sixty 16 ms 2000V pulses were applied between LL 1 and LL 2 without realizing spark gap A 1 was missing with out damage. During normal operation the NSME L 1 flux density is a few hundred gauss. The 75 u material from the graph of Flux Density v. Magnetizing Force (FIG. 15) will accept flux densities at least 50 times greater with out limitation. This is an example of the magnetic elements ability to perform well at flux densities many times greater than prior art. Elements L 1 and L 2 will block differential or common mode line transients. In the event of a very large or long duration line to neutral transient, spark gap A 1 will clamp the voltage to a safe level of about 400V. The NSME L 1 and L 2 have the added benefit of reducing conducted noise generated by the converter. FIG. 21 Alternate Lightning Input Protection Sub-circuit LLA FIG. 21 is a schematic showing the inventions alternate lightning protection sub-circuit for an AC line connected converter. The protection circuit comprises a fuse F 1 , Spark gap A 1 , capacitors C 1 , C 60 and C 61 and NSME L 3 . FIG. 21 Table Element Value/part number L3 750 uH C61 0.01 uF C60 0.01 uF F1 10A A1 400 V Spark Gap C1 0.1 uF D20 1000 V/25A D21 1000 V/25A D22 1000 V/25A D23 1000 V/25A C2 1.8 uf High side of AC line is connected to node LL 2 , fuse F 1 the load side of the fuse is connected to NSME L 3 and to capacitor C 61 . The load side L 3 is connected to spark gap A 1 and the cathode of diode D 20 and anode of D 22 forming node LL 5 . The low side of AC line is connected to node LL 6 , capacitor C 60 , spark gap A 1 , and the cathode of diode D 21 and anode of D 23 . The anodes of diodes D 20 and D 21 are connected to Capacitor C 1 . Capacitor C 1 is connected to earth ground. C 1 attenuates radiated noise or EMI generated by the converter. The cathode of diodes D 22 and D 23 are connected to Capacitor C 2 . Capacitor C 2 decouples high frequency harmonic currents from the line. Capacitors C 1 , C 61 and C 60 are connected to earth ground node LL 0 . The use of non-saturating magnetics allows the input magnetic element to absorb very large voltages and currents commonly generated on the AC line by lightning. A transient on the AC line will be limited by capacitors C 60 and C 61 and blocked by the non-saturating magnetic L 3 . In the event of a very large or long duration line to neutral transient, magnetic element L 3 will allow the voltage to rise across spark gap A 1 , the spark gap will clamp the voltage to a safe level protecting the rectifier diodes D 20 -D 23 . The NSME L 3 has the added benefit of reducing conducted noise generated by the converter. C 1 connected to the ground plane is effective in attenuating conducted and radiated EMI. FIG. 22 AC Line Rectifier Sub-circuit BR FIG. 22 is a schematic showing the inventions AC line rectifier. The rectifier sub-circuit BR 1 comprises diodes D 20 , D 21 , D 22 and D 23 and capacitor C 2 . FIG. 22 Table Element Value/part number D20 1000 V/25A D21 1000 V/25A D22 1000 V/25A D23 1000 V/25A C2 1.8 uf An AC or DC signal from the input filter is connected to bridge rectifier to nodes BR 1 and BR 2 . Node BR 1 connects diode D 22 anode to D 20 cathode. Node BR 2 connects diode D 23 anode to D 21 cathode. Node BR+ connects diode D 22 cathode to D 23 cathode. Node BR− connects diode D 20 anode to D 21 anode. The common input frequencies of DC to 440 Hz may be extended beyond this range with component selection. Capacitor C 2 is selected to improve power factor for a particular operating frequency and to de-couple switching currents from the line. Diodes are selected to reliably block the expected line voltage and current demands of the next converter stage. FIG. 23 Controller Sub-circuit PFA FIG. 23 is the inventions AC to DC controller sub-circuit. Sub-circuit PFA consists of resistors R 313 and R 316 , capacitors C 308 , and C 313 and PWM controller IC U 1 A. FIG. 23 Table Element Value/part number C311 0.1 uf C308 .01 uf R313 15k ohms R316 15k ohms C313 4700 pf R308 25k ohms U1A MIC38C43 (Micrel) Control element U 1 A connects to a circuit with the following series connections: from pin 1 to feedback node/pin PF 1 then to capacitor C 308 then to the pin 2 node of U 1 A. Internal 5.1-volt reference U 1 A pin 8 or node PFA 2 through resistor R 308 to the pin 4 node. U 1 A pin 4 is connected through capacitor C 313 to return node BR−. This arrangement allows the PFC output to be pulse width modulated with application of voltage to PF 1 . External feedback current applied to U 1 A pin 1 and node PF 1 . Node PFVC is connected to resistor R 313 to pin 3 of U 1 A. Resistor R 316 is connected to pin 3 then to return node BR−. Power pin 7 is connected to node PFA+. Control element switch drive U 1 A pin 6 is connected to node PFCLK. Return ground node of U 1 A pin 5 is connected to return node BR−. In the event of a component failure in the primary feed networks such as IPFFB (FIG. 40 ), FBA (FIG. 40 A), IFB (FIG. 40B) and FB 1 (FIG. 41 ). The output voltage of the boost stage may rapidly increase to destructive levels. Fast over voltage feedback networks IOVFB (FIG. 40C) or OVP 2 (FIG. 42B) increases the current into PF 1 thereby limiting the output voltage to a safe level. In addition latching type over voltage protection networks such as OVP (FIG. 42 ), OVP 1 (FIG. 42A) and OVP 2 (FIG. 42B) maybe used. The latching type kills power to the control circuit thereby stopping the boost action. The latching type networks require power to be cycled to the converter to reset the latch. IFB Input node PFVC is connected to resistor R 313 to internal zero crossing detector connected to pin 3 and through R 316 to return node BR−. PFVC is connected to a magnetic element winding referenced to BR−. A new conduction cycle is started each time the bias in the magnetic element goes to zero. Power factor corrected is realized by chopping the input at a high frequency. The average pulse width decreases at higher line voltage and increases at lower voltage for a given load. Frequency is lower at line peaks and higher around zero crossings. In this way the converter operates with a high input power factor. FIG. 24 Controller With Power Factor Corrected Sub-circuit PFB FIG. 24 is the alternate power factor controller sub-circuit. Sub-circuit PFB consists of resistors R 313 , R 339 , R 314 , R 315 , R 328 , R 340 , R 341 and R 346 , diode D 308 , capacitors C 310 , C 318 , C 338 , C 340 , C 341 and C 342 , transistor Q 305 , and control element IC U 1 B. FIG. 24 Table Element Value/part number Q305 BCX70KCT R339 432k ohms C338 0.22 uf C318 0.22 uf R314 2.2 MEG ohms R315 715k ohms C341 0.33 uf C342 0.01 uf C340 0.001 uf R328 1 MEG ohms R346 7.15k ohms D308 10BQ040 R340 449k ohms R313 22k ohms U1B MC34262 (Motorola) R341 499k ohms Control element U 1 B connects to a circuit with the following series connections: from pin 1 to node/pin PF 1 to capacitor C 338 in series with resistor R 339 , and then to the pin 2 node of U 1 B. Pin 1 is the input to an internal error amplifier and connection to external feedback networks. (See FIG. 40, 40 A, 40 B, 40 C and 41 ) Increasing the voltage on pin 1 decreases the pulse width of the PFCLK node pin 7 . Resistor R 328 is connected to the fullwave rectified AC line haversine voltage on node BR+ then to U 1 B pin 3 and then to resistor R 346 in parallel with capacitor C 342 to return node BR−. Node PFSC connects to series resistors [R 341 +R 340 ] which are connected to U 1 B pin 4 then to diode D 308 in parallel with capacitor C 340 to return node BR−. Power to PFC controller is applied to node PFB+ and to U 1 B pin 8 . Output clock node PFCLK is connected to U 1 B pin 7 , to external buffer sub-circuit AMP (FIG. 29 ). Transistor Q 305 collector is connected to the pin 2 node of U 1 B. The base is connected in series through resistor R 314 to capacitor C 318 , then to the pin 2 node of U 1 B. The base is also connected to [C 310 ||R 315 ], then to return node BR−. Emitter of Q 305 is connected to return node BR−. Transistor Q 305 provides a soft start compensation ramp to the controller error amp reducing the stress and DC overshoot in the converter at power up. Capacitor C 341 is connected from U 1 pin 2 to return node BR−. U 1 B pin 1 is connected to pin PF 1 , capacitor C 338 in series with resistor R 339 to transistor Q 305 collector and to U 1 pin 2 . Current switched by PFC power switch Q 1 (FIG. 4 & 3) is sensed by R 26 (see FIG. 4 ). Series resistors [R 340 +R 341 ] to U 1 B pin 4 connect voltage developed across R 26 . This voltage is compared to an internal 1.5-volt reference, comparator. output turns off the switch drive pin 7 of U 1 B during times of high current that occur during startup or during very high load or low line conditions. Capacitor C 340 is connected between U 1 pin 4 to return node BR− filter high frequency components. Schottky diode D 308 connected between U 1 pin 4 to return node BR− protects the controller (U 1 pin 4 ) substrate from negative current injection. Maximum switch current value is set by R 26 over currents are automatically limited in each cycle by the PFC controller. The rectified fullwave haversine at pin 3 of U 1 B is multiplied by the error voltage on pin 2 . The product is compared to the magnetic element/switch current measured by R 26 on pin 4 . Gate drive on pin 7 turns off when the sensed magnetic element current increases to the current comparator level. This action has the effect of modulating the switch Q 1 “on” time tracking the AC line voltage. External feedback networks are connected to node PF 1 . In the event of a component failure in the primary feed network such as IPFFB (FIG. 40 ), FBA (FIG. 40 A), IFB (FIG. 40B) and FBI (FIG. 41 ). The output voltage of the boost stage may rapidly increase to destructive levels. Fast over voltage feedback networks IOVFB (FIG. 40C) or OVP 2 (FIG. 42B) increases the current into PF 1 thereby limiting the output voltage to a safe level. In addition latching type over voltage protection networks such as OVP (FIG. 42 ), OVP 1 (FIG. 42A) and OVP 2 (FIG. 42B) maybe used. The latching type removes power to the control circuit thereby stopping the boost action. The latching type networks require power to be cycled to the converter to reset the latch. Modulating the voltage at PF 1 changes the duty cycle of the PFC and the final output voltage. In this way the PFC may be used as a pre-regulator to additional output stages. FIG. 25 Output Rectifier and Filter Sub-circuit OUTA FIG. 25 is a schematic of a full wave rectified output stage and filter sub-circuit OUTA. The rectifier stage 10 consists of diodes D 80 and D 90 . The filter consists of resistor R 21 , magnetic element L 30 and capacitors C 26 , C 27 , C 28 , C 29 , C 30 , C 31 and C 32 . FIG. 25 Table Element Value/part number D80 40CTQ150 D90 40CTQ150 R21 100 ohms C26 500 pf C27 200 pf C28 0.1 uf C29 10,000 uf C30 10,000 uf C31 0.1 uf C32 200 pf L30 10 uh Input node/pin C 7 B is connected to the high side of external center-tapped magnetic element secondary winding. Node C 7 B connects to anode of diode D 8 and to capacitors C 26 and C 27 in the following arrangement. Capacitor C 27 is connected across diode D 80 , capacitor C 26 is connected in series to R 21 . Input node/pin C 8 B is connected to the low side of external center-tapped magnetic element secondary winding. Pin C 8 B is connected to anode of diode D 9 and to resistor R 21 , capacitor C 32 is connected across diode D 90 . Capacitors C 27 and C 32 is a small value to reduce high frequency noise generated by rapid switching the high speed rectifier D 80 and D 90 respectively. Capacitor C 26 and resistor R 21 are used to further dissipate high frequency energy. Anode of diodes D 80 and D 90 is connected to parallel capacitors C 281 C 29 and NSME L 30 . Capacitors C 28 and C 31 are solid dielectric types selected for low impedance to high frequency signals. Capacitors C 29 and C 30 are larger polarized types selected for low impedance at low frequencies and for energy storage. Magnetic element L 3 is connected to diode D 8 cathode the second terminal of L 30 is connected to parallel capacitors C 31 and C 30 and pin OUT+. Node OUT+ is the output positive and is connected to external feedback sense line to isolated feedback network. The other side of parallel capacitors [C 28 ||C 29 ||C 30 ||C 31 ] is connected to pin OUT− and the center-tap of the magnetic element secondary forming the return node. The combination of capacitors [C 28 ||C 29 ], L 30 and capacitors [C 30 ||C 31 ] form a low pass pi type filter. Sub-circuit OUTA performs efficient fullwave rectification and filtering. FIG. 25A Output Rectifier Sub-circuit OUTB FIG. 25A is a schematic of a full wave rectified output stage. The rectifier stage consists of diodes D 80 and D 90 and capacitors C 931 and C 928 . FIG. 25A Table Element Value/part number D80 40CTQ150 D90 40CTQ150 C928   .01 uf C931 10,000 uf Input node/pin C 7 B is connected to high side of external center-tapped magnetic element secondary winding. Node C 7 B connects to anode of diode D 80 . Input node/pin C 8 B is connected to low side of external center-tapped magnetic element secondary winding is connected to anode of diode D 90 . Node OUT− is the negative output and return line to the external isolated feedback network and load not shown. Cathodes of diodes D 80 and D 90 are connected to parallel capacitors C 931 and C 928 . Capacitor C 928 is a solid dielectric type selected for low impedance to high frequency signals. Capacitor C 931 is a larger polarized selected for low impedance to low frequency signals and for energy storage. Node OUT+ is the output positive and is connected to external feedback sense line to isolated feedback network. The other side of parallel capacitors C 928 ||C 931 is connected to the center-tap of the magnetic element secondary forming the node OUT−. The use of the NSME for the push-pull magnetic element requires only minimal filtering after the rectifiers. FIG. 25B AC Rectifier and Filter Sub-circuit OUTB FIG. 25B is a schematic diagram of an alternate final output rectifier and filter sub-circuit OUTB. The rectifier sub-circuit OUTB comprises diodes D 40 , D 41 , D 42 and D 43 and capacitor C 931 and C 928 . FIG. 25B Table Element Value/part number D40 40CTQ150 D41 40CTQ150 D42 40CTQ150 D43 40CTQ150 C928   .01 uf C931 10,000 uf An AC or DC signal is connected to nodes C 7 B and C 8 b . Node C 7 B connects diode D 41 anode to D 40 cathode. Node C 8 b connects diode D 42 anode to D 43 cathode. Node OUT+ connects diode D 42 cathode to D 43 cathode. Node OUT− connects diode D 40 anode to D 43 anode. Diodes are selected to reliably block the expected line voltage and current demands of the load. For low voltage outputs, Schottky type diodes are used due to their low forward voltage drop. Higher voltages would use high-speed silicon diodes due to their ability to withstand high peak inverse voltage (PIV). The use of the NSME for the push-pull magnetic element requires only minimal filtering after the rectifiers. Capacitor C 928 is shown schematically as a single device. Capacitor C 931 is a larger polarized selected for low impedance to low frequency signals and for energy storage a typical value may be 200 uF. To increase the capacitance or reduces the output impedance multiple capacitors may be used. C 931 is a solid dielectric type and is selected for it's low impedance to high frequencies. As is selected to reduces noise for a particular operating frequency and power level. Capacitor C 928 is selected for the operating frequency and power level. Sub-circuit OUTB performs AC to DC rectification and filtering at slightly lower efficiency due to the extra junctions. FIG. 26 Floating 18_Volt DC Control Power Sub-circuit CP Sub-circuit CP consists of diodes D 501 , D 502 and D 503 , resistor R 507 , regulator Q 504 , and capacitors C 503 , C 504 , C 505 , C 506 , and C 507 . FIG. 26 Table Element Value/part number C503 .33 uF C504 100 uF D501 MURS120T3 C505 .33 uf Q504 LM7818A C508 100 uf C507 100 uf D503 MURS130T3 D502 MURS120T3 Node CT 1 A connects to anode of D 503 and to the upper external center tapped secondary winding. Node CT 2 A connects to anode of D 502 and to the lower external center tapped secondary winding. Node CT 0 connects to the external winding center tap. Node CT 0 is also the return line and it connects to Q 504 pin 2 , and capacitors C 503 , C 504 , C 505 , C 506 , and C 507 . The cathode of each of diodes D 502 and D 503 is connected to resistor R 507 . R 507 is then connected to the pin 1 (input) node of voltage regulator Q 504 . Voltage regulator Q 504 Pin 3 is the 18vdc regulated DC output is connected to the anode of blocking diode D 501 . Three-pin voltage regulator Q 504 is of the type LM7818 a common device made by a number of manufacturers. Capacitors C 503 , C 505 , C 506 are 0.1 uF solid dielectric type and are used to filter high frequency ripple and to prevent Q 504 from oscillating. The junction of C 503 , C 504 and D 501 cathode is the output node CP 1 +. Isolated 18-volt DC is available between nodes CT 0 and CP+. Used for regulator circuits and output switch drive during normal operation. FIG. 27 second Floating 18_Volt DC push-pull control power sub-circuit CPA. Sub-circuit CPA consists of diodes D 601 , D 602 and D 603 , resistor R 607 , regulator B 604 and capacitors C 603 , C 604 , C 605 , C 606 , C 607 and C 608 . FIG. 27 Table Element Value/part number C603 .33 uF C604 100 uF D601 MURS12OT3 C605 .33 uF Q604 LM7818A C608 100 uF C607 .22 uF R607 7.5 ohms D603 MURS120T3 D602 MURS120T3 Node CT 1 B connects to anode of D 603 and to the upper external center tapped secondary winding. Node CT 2 B connects to anode of D 602 and to the lower external center tapped secondary winding. Node CT 20 connects to the external winding center tap. Node CT 0 is also the return line and it connects to Q 604 pin 2 , and capacitors C 603 , C 604 , C 605 , C 606 , and C 607 . The cathode of each of diodes D 602 and D 603 is connected to resistor R 607 . R 607 is then connected to the pin 1 (input) node of voltage regulator Q 604 . Voltage regulator Q 604 Pin 3 is the 18 vdc regulated DC output and is connected to the anode of blocking diode D 601 . Capacitors C 603 , C 605 , C 606 are solid dielectric type and are used to filter high frequency ripple and to prevent Q 604 from oscillating. The junction of C 603 , C 604 and D 601 cathode is the output node CP 1 +. Isolated 18-volt DC is available between nodes CT 20 and CP 2 +. To be used for regulator circuits and output switch drive during normal operation. FIG. 28 Over Temperature Protection Sub-circuit OTP FIG. 28 is the main switch over temperature protection sub-circuit OTP. The sub-circuit OTP comprises thermal switch and resistors R 711 and R 712 . FIG. 28 Table Element Value/part number THS1 67F105 (105C) R711 20 ohms R712 20 ohms Gate drive power is applied to input node GAP and to thermal switch THS 1 . Maximum FET gate voltage requires the input power voltage be less than 20 volts, the voltage selected was 18 volts. The other side of THS 1 is connected to parallel resistors [R 711 ||R 712 ]. A single resistor may represent the resistors. The figure depicts the surface mount arrangement. The other side of [R 711 ||R 712 ] connects to output node TS+. Normally closed thermal switch TS 1 is in contact with main switch transistor Q 1 . In the event of temperature greater than 105C THS 1 opens, thus removing power to the buffer sub-circuit AMP 1 (FIG. 29) causing switch Q 1 to default to a blocking state protecting the boost switch should the optional cooling fan fail or the circuit reach high temperatures. In this instant invention the speed up buffer AMP (FIG. 29) non-saturating magnetics (FIG. 18, 18 A and 19 ) allows the main switch and to run cooler than prior art for a given power level. When switch temperature returns to normal range THS 1 will close, allowing the PFC to resume normal operation. Under normal load and ambient temperatures the thermal switch THS 1 should never open. FIG. 29 PFC Buffer Circuit Sub-circuit AMP, AMP 1 , AMP 2 , AMP 3 Switch drive command from PFCLK (FIG. 23 and 24) or PWFM (FIG. 33) control elements are connected to a gate buffer circuit. The sub-circuit AMP is comprised of power FET Q 702 , Darlington pair Q 703 , capacitors C 709 and C 715 , and resistors R 710 and R 725 . FIG. 29 Table Element Value/part number C715 1000 pf C709 33 uF Q702 NOS355NCT Q703 FZT705CT R710 0 ohms R725 22.1k ohms DC Power is applied to node GAT+ to transistor Q 702 drain and to capacitor C 709 , which goes to ground. Maximum gate voltage requires the input power voltage must be less than 20 volts, 18-volts was selected. Input node GA 1 is connected to the gate of FET Q 702 is connected to the base of BJT 1 of the Darlington pair Q 703 and to capacitor C 715 . C 715 is connected across the Darlington pair from the base, pin 1 , to the collectors, pins 2 and 4 , Q 703 collector node is also connected to ground. The emitter of BJT 2 is connected to the gate of FET Q 1 . The source of FET Q 702 is connected through small optional series resistor R 710 to the gate of the output switch or node GA 2 . Some power FET's under certain load may tend to oscillate when driven from a low impedance source such as this buffer. A small resistance of approximately 2 ohms or less may be required with out significant slowing of the switch. In most cases R 710 is replaced with a zero ohm jumper. Resistor R 725 is connected from node GA 0 and source of Q 702 . The input switching signal to node GAP is in range of 20 kHz to 600 kHz. Very fast “on” times are realized by proving a low impedance to rapidly charge the output switch gate connected to node GA 2 . Capacitor C 709 provides additional current when Q 702 switches on. Transistor Q 703 provides low impedance to rapidly remove the charge from the gate greatly reducing the “off” time. This particular topology provides output switch rise times on the order of 10 ns, as compared to the industry standard rise time of 250 ns. The corresponding fall time is <10 nS, again as compared to an industry fall time of 200-300 ns (See FIG. 13 and 14 ). In the event the converter is operated at very high ambient temperatures a thermal switch may be placed in series with input power pin GA+. This allows the switch transistor to be gracefully disabled. Sub-circuit AMP greatly reduces switching losses allowing converter operation in some cases with out the common prior art forced air-cooling. FIG. 30 Snubber Sub-circuit SN FIG. 30 is a schematic diagram of a snubber sub-circuit of the invention. The snubber sub-circuit SN is comprised of diodes D 804 and D 805 and resistors R 800 , R 817 , R 818 , and capacitors C 814 and C 819 . Table FIG. 30 Value/ Element part number R800 12 ohms R817 1 mohm R818 1 mohm C814 33 pF C819 560 pF D805 MUR160 Node SNL 2 connects to the drain terminal of the external output switch and to flyback side of the inductive load. Input node SNL 2 connects to R 800 in series with capacitor C 819 to node SNOUT. Diode D 805 anode is connected to node SNL 2 with resistors [R 817 ||R 818 ] in parallel with D 805 . Resistors R 817 and R 818 may be combined to a single resistor. The cathode of D 805 is connected to capacitor C 814 that connects to node/pin SNL 1 . Node SNL 1 connects to the supply side of external load magnetic element. The other leg of external magnetic element is connected to the anode of D 805 and the anode side of external flyback diode D 4 . The one MEG ohm resistors R 817 and R 818 bleed the charge from C 814 resetting it for the next cycle. Capacitor C 819 and resistor R 800 captures the high frequency event from the transition of external flyback diode D 4 and moves part of the energy into the external holdup capacitor connected to node SNOUT. Since external flyback diode D 4 and D 805 isolate the drain of the output switch, faster switching occurs because the output switch does not have to slew the extra capacitance of a typical drain/source connected snubber circuit. Note that this circuit does not attempt to absorb the flyback in large RC networks that convert useful energy to losses. Nor does it attempt to stuff the flyback to ground, adding capacitance and slowing the output switch and increasing switching losses. This sub-circuit is used with it's mirror SNB (FIG. 32) across the external push-pull switches. This design returns the some of the flyback energy back to the input supply or output load. The “snubbering” action slows the rise of the flyback giving time for the external flyback diode to start conduction. The circuit efficiently manages high frequency flyback pulses. FIG. 30A Diode Snubber Sub-circuit DSN FIG. 30A is a schematic diagram of a diode snubber sub-circuit of the invention. The snubber sub-circuit DSN is comprised of diodes D 51 , D 52 , D 53 , D 54 and D 55 and capacitors C 51 , C 52 , C 53 , C 54 and C 55 . FIG. 30A Table Element Value/part number C51 220 pf 100 v C52 220 pf 100 v C53 220 pf 100 v C54 220 pf 100 v C55 220 pf 100 v D51 Schottky 1-2 ns 100 v SMBSR1010MSCT D52 Schottky 1-2 ns 100 v SMBSR1010MSCT D53 Schottky 1-2 ns 100 v SMBSR1010MSCT D54 Schottky 1-2 ns 100 v SMBSR1010MSCT D55 Schottky 1-2 ns 100 v SMBSR1010MSCT Pin SNL 2 is connected to the anode of D 51 the cathode of D 51 is connected to the anode of D 52 the cathode of D 52 is connected to the anode of D 53 the cathode of D 53 is connected to the anode of D 54 the cathode of D 54 is connected to the anode of D 55 the cathode of D 55 is connected to pin SNOUT. Capacitors are connected across each diode forming a series parallel combination of [D 51 ||C 51 ]+[D 52 ||C 52 ]+[D 53 ||C 53 ]+[D 54 ||C 54 ]+[D 55 ||C 55 ]. Node SNL 2 connects to the drain terminal of the external output switch and to flyback side of the inductive load. The external fly-back rectifier diode D 4 (FIG. 1, 3 and 4 ) anode is connected to node SNL 2 . Node SNOUT connects to the storage capacitors [C 16 ||C 17 ] (FIG. 1, 3 and 4 ) and to the cathode of the flyback diode D 4 . External diode D 4 in parallel with DSN forms a hybrid diode. The Schottky diode has the desirable characteristics of fast recovery time (less than 6 nanoseconds (6* 10{circumflex over ( )}−9)) and low forward voltage drop (0.4-0.9 Volts) at high currents. The Schottky diode suffers from limited reverse blocking voltage currently 100 V maximum. Each diode will block 100V; the parallel capacitors distribute the reverse voltage equally across the diode string. As the reverse junction capacitance of each diode is less than 10 pf much smaller than the parallel capacitor. Thus the reverse voltage is nearly equally divided across the diodes. To guarantee even voltage division 5% or better capacitor matching is required. High precision is common and inexpensive for small capacitors. Different blocking voltages may be achieved by adjusting the number of diode/capacitor pairs. By way of example not as a limitation 500V was selected. The main fly-back rectifier diode D 4 will block high voltages but suffers from long reverse recovery time 50-500 nanoseconds is common in fast recovery diodes. What is needed is a diode with low voltage drop, high blocking voltage and very short recovery time. The snubber DSN in parallel with the main fly-back rectifier comes very close to that ideal diode. The total blocking voltage is achieved by the adding the individual diode blocking voltages. The recovery time is determined by the slowest diode in the string often less than 5 nanoseconds. The low forward voltage drop is achieved when the slower main rectifier begins conduction. Low capacitance is also realized, as the capacitance is ⅕ of the individual capacitors. This hybrid diode begins rectification immediately after the main switch stops conduction and the non-saturating magnetic begins releasing its energy. This effectively limits the high voltage flyback over shoot to less than 40-70 volts. This keeps the switch well inside it's safe operating area (SOA) allowing the switch to be run at higher voltages for higher output power and additional efficiency gain, or to use a less expensive lower voltage switch while keeping the same voltage margins. Since external flyback diode D 4 and D 805 isolate the drain of output switch, faster switching occurs because the output switch does not have to slew the extra capacitance of the typical snubber circuit. Note that this circuit does not attempt to absorb the flyback in large RC networks that generate additional heat. Nor does it attempt to stuff the flyback to ground, adding capacitance and slowing the output switch, increasing switching losses. Sub-circuit DSN may be used in parallel with any slower rectifier such as flyback diode D 4 to assist the main rectifier. This providing additional protection to the switch and rectifying the portion of the flyback pulse before the main rectifier begins condition. That high frequency energy ends up as heat or radiated noise. FIG. 31 Snubber Sub-circuit SNA FIG. 31 is a schematic diagram of a snubber sub-circuit of the invention. The snubber sub-circuit SNA is comprised of resistor R 810 and R 811 and capacitors C 820 and C 821 . FIG. 31 Table Element Value/part number R810 500 pF C811 330 pF C820  12 ohm C821  10 ohm Node SNA 1 connects to series resistor R 810 to capacitor C 820 to node SNA 2 then to capacitor C 821 and series resistor R 811 to node SNA 3 . Node SNA 1 connects to the external magnetic element center tap. Node SNA 2 connects to the drain terminal of the external output switch and to flyback side of the inductive load. Node SNA 3 connects to the source terminal of the external output switch. Resistor R 810 and C 820 attempt to absorb part of the flyback to reduce voltage transients across the switch. Part of the flyback is returned to ground by C 821 . This sub-circuit is used with itís mirror SNA (FIG. 31) across the external push-pull switches. The “snubbering” action slows the rise of the flyback giving time for the external rectifier diodes D 8 and D 9 of FIG. 25 or 25 A to start conduction. The circuit efficiently manages high frequency flyback pulses. FIG. 32 Snubber Sub-circuit SNB FIG. 32 is a schematic diagram of a snubber sub-circuit of the invention. The snubber sub-circuit SNB comprises resistor R 820 and R 821 and capacitors C 840 and C 841 . FIG. 32 Table Element Value/part number C840 500 pF C841 330 pF R820  12 ohm R821  10 ohm Node SNB 1 connects to series resistor R 820 to capacitor C 820 to node SNA 2 to capacitor C 841 and to series resistor R 821 to node SNB 3 . Node SNB 1 connects to the external magnetic element center tap. Node SNB 2 connects to the drain terminal of the external output switch and to flyback side of the inductive load. Node SNB 3 connects to the source terminal of the external output switch. Resistor R 820 and C 840 attempt to absorb part of the high frequency flyback to reduce voltage transients across the switch. C 841 and R 821 return part of the flyback to ground. The “snubbering” action slows the rise of the flyback giving time for the external rectifier diodes D 8 and D 9 of FIG. 25 or 25 A to start conduction. The circuit efficiently manages high frequency flyback pulses. FIG. 33 Pulse/Frequency Modulator PWFM FIG. 33 is the inventions PWM (pulse width modulator) and FM (frequency modulator) sub-circuit. Sub-circuit PWFM consists of resistors R 401 , R 402 , R 403 , and R 404 capacitors C 401 , C 402 , C 403 , C 404 , C 405 and C 406 , controller IC U 400 and diode D 401 . FIG. 33 Table Element Value/part number R404 50k ohms C406 100 uf C401 0.22 uF C403 0.01 uF C405 2200 pF C404 470 pF C402 0.22 uF R403 50k ohms D401 RLS139(low leakage) R401 2.2 MEG ohms R402 150k ohms U400 MIC38C43 Control element U 400 connects to a circuit with the following series connections: from pin 1 to feedback pin PW 1 then to the wiper of adjustable resistor R 404 to return node PWFM 0 . Resistor R 404 may be replaced with two fixed resistors. Capacitor C 403 is connected from pin 2 to pin 1 . Capacitor C 403 is used to filter the error amp output. The upper half of resistor R 404 is connected to node REF 1 pin 8 the 5.0-Volt internal reference. Internal 5.0-volt reference U 400 pin 8 or Node REF 1 is connected to the upper half of resistor R 403 and through capacitor C 402 to return node PWFM 0 . The reference provides current to external feed back networks. Wiper of R 403 connects to node FM 1 to pin 4 , through R 402 to pin 3 , and through C 404 to return node PWFM 0 . Resistor R 403 may be replaced with two fixed resistors. Pulse width timing capacitor C 404 connects pin 3 to return node PWFM 0 . Low leakage diode D 401 anode is connected to pin 3 cathode to output pin 6 node CLK. Resistor R 404 sets the nominal pulse width of output pin 6 node CLK. The pulse width can be adjusted from 0 (off) to 95%. Resistor R 403 and C 404 determine the nominal operating frequency. With application of power 20-volts between Nodes PWFM+ and PWFM 0 controller U 400 generates an internal 5.0 reference voltage to pin 7 node REF 1 . Output pin 6 node CLK is set high approximately 20-volts (see oscillograph trace G 6 segment 60 FIG. 34 ). C 404 starts to charge through R 401 until the voltage across C 404 at pin 3 reaches the comparator level (see oscillograph trace G 1 segment 61 FIG. 34) at resetting the pin 6 low (see oscillograph trace G 6 segment 62 FIG. 34 ). Capacitor C 404 rapidly discharges though D 401 (see oscillograph trace G 1 segment 63 FIG. 34 ). Pin 3 remains 0.6-volts above PWFM 0 node during the period pin 6 is low (see oscillograph trace G 1 segment 64 FIG. 34 ). On the rising edge of pin 6 capacitor C 405 begins to rapidly charge until the voltage in pin 4 reaches the internal comparator level (see oscillograph trace G 4 segment 65 FIG. 34 ). The comparator triggers internal transistor to rapidly discharge C 404 (see oscillograph trace G 4 segment 66 FIG. 34 ). The cycle repeats with output pin 6 being set high. External feedback current applied to U 400 pin 1 and node PW 1 (see oscillograph trace G 1 segment FIG. 34) follows the actual output voltage. Oscillograph trace G 1 segment 67 (FIG. 34) is the period when the output switch conducting storing energy in the NSME. Oscillograph trace G 1 segment 68 (FIG. 34) is the period when the output switch is off allowing storing energy in the NSME to be transferred to the storage capacitor. Application of external current source or feed back network to pin 1 or node PW 1 allows the pulse width to be modulated. Removing current from PW 1 lowers the comparator level causing the comparator to trigger at lower voltages across C 404 reducing the pulse width. Introducing current into node PW 1 increases pulse width from nominal to maximum of 95%. Resistor R 404 and C 404 determine the nominal pulse width. This design allows the CLK output to be pulse width modulated. Application of external feed back network to pin 4 or node FW 1 allows the frequency to be modulated. Removing current from FW 1 slows the charging of C 405 . Longer charging time lowers the frequency from the nominal setting. This arrangement allows the CLK output to frequency modulated. When used with a resonant controller, R 403 and C 405 determine the nominal frequency typically equal to the tank resonant frequency. The external feedback is configured to lower the frequency from nominal (maximum output) to zero frequency “off”. When used as a pulse-width controller the nominal is set to maximum pulse width of about 90% feedback reduces the pulse-width. Sub-circuit PWFM may be simultaneously frequency and pulse width modulated. This configuration and mode of operation is unique to this instant invention. Feeding back of the output to the error amplifier is a unique mode of operation for control element U 400 . Sub-circuit PWFM combines large dynamic range, precise control and fast response. FIG. 34 Oscillograph traces of the PWFM (FIG. 33) controller in the pulse-width modulation mode. FIG. 35 Oscillograph trace of the TCTP (FIG. 8) resonant converter primary voltage. FIG. 35 is an oscillograph trace of the voltage developed across capacitor C 10 (FIG. 8 ). In this embodiment the supply VBAT was only 18-volts. The primary 100 (FIG. 18) inductance 203 uH was achieved by 55 turns on a 26 u 2.28 oz. KoolMu magnetic element 101 . The secondary winding 103 (FIG. 18) is 15 turns on core 101 . A 5.5-watt load is connected to winding 103 . The NSME primary 100 (FIG. 18) developed an excitation voltage of 229 volts peak more than 10 times VBAT. Tank converters TCTP and TCSSC (FIG. 7) take advantage of the desirable properties of the non-saturating magnetic to develop large flux biases. The useful large flux may harvested into useful power by addition of “flux nets” windings to the magnetic element. FIG. 36 Regulated 18_Volt DC Control Power Sub-circuit REG Sub-circuit REG consists of resistor R 517 , regulator Q 514 and capacitors C 514 , C 515 , C 516 , C 518 , and C 517 . FIG. 36 Table Element Value/part number Q514 LM7818 C515 0.1 uF C517 0.1 uF C514  10 uF C518  10 uF Pin REG 0 connects to the external power source return. Node REG 0 is also the return line it connects to Q 514 pin 2 , and capacitors C 518 , C 514 , C 515 , and C 517 . Resistor R 517 is connected to the pin 1 (input) node of voltage regulator Q 514 and to input pin RIN+. Voltage regulator Q 514 Pin 3 is the 18vdc regulated DC output is connected to the capacitors C 515 , C 514 and output pin 18 V. Capacitors C 515 , C 517 are solid dielectric type is used to filter high frequency ripple and to prevent Q 514 from oscillating. Sub-circuit REG provides regulated power for control circuits and output switch buffer AMP (FIG. 29 ). FIG. 37 is a schematic for a non-isolated high side switch buck converter sub-circuit HSBK. FIG. 37 is a non-isolated high side switch buck converter sub-circuit HSBK. This converter topology consists of a non-isolated high efficiency buck stage, which provides regulated power to an efficient push-pull isolation stage. Sub-circuit HSBK consists of diode D 8 , capacitor C 8 , FET transistor Q 31 , sub-circuit TCTP (FIG. 8 ), sub-circuit BL 1 (FIG. 18 B), sub-circuit IFB (FIG. 40 B), sub-circuit AMP (FIG. 29) and sub-circuit PWFM (FIG. 33 ). FIG. 37 Table Element Value/part number C68 250 uf D68 MUR820 Q31 IRF540N External power source VBAT connects to pins DCIN+ and DCIN−. Pin DCIN+ connects to transistor Q 31 source, sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , and sub-circuit IFB pin FBE, sub-circuit TCTP pins DCIN+ and B−. Regulated 18-volt output from sub-circuit TCTP pin B+ connects to sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. This provides the positive gate drive relative to the source of Q 31 . Power source VBAT return is connected to pin DCIN−, sub-circuit TCTP pin DCIN−, diode D 68 anode, capacitor C 68 , RLOAD, sub-circuit IFB pin OUT−, output pin B− and ground/return node GND. Sub-circuit PWFM is designed for adjustable pulse-width operation from 0 to 90%, maximum pulse width occurs with no feedback current to pin PW 1 . Increasing the feedback current reduces the pulse-width and output voltage from converter HSBK. Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP. The output of sub-circuit AMP pin GA 2 is connected to the gate of Q 31 . The drain of Q 31 is connected to sub-circuit BL 1 pin P 1 B and the cathode of D 68 . Pin P 1 A of sub-circuit BL 1 is connected to capacitor C 8 , sub-circuit IFB pin OUT− and RLOAD. With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 31 . Switch Q 31 conducts charging capacitor C 68 through NSME BL 1 from source VBAT and storing energy in BL 1 . Feedback output pin FBC from sub-circuit IFB is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . As the output voltage reaches the designed level sub-circuit IFB removes current from PW 1 commanding PWFM to reduce the pulse-width or on time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PWFM switches output pin CLK low turning off Q 31 stopping the current into BL 1 . The stored energy is released from NSME BL 1 into the now forward biased diode D 68 charging capacitor C 68 . By modulating the on time of switch Q 31 the converter “bucks” applied voltage and efficiently regulates to a lower voltage. Regulated voltage is developed across Nodes B− and B+. Sub-circuit IFB provides the isolated feedback voltage to the sub-circuit PWFM. When sub-circuit IFB senses the converter output (nodes B+ and B−) is at the designed voltage more current is conducted by the phototransistor. Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus reducing the converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 31 is removed stopping all buck activity capacitor C 68 discharges through RLOAD. Input current from VBAT is sinusoidal making the converter very quiet. As such the switch Q 31 is not exposed to large current spikes common to saturating magnetic prior art. Thus placing less stress on the switches thereby increasing the MTBF. Sub-circuit HSBK takes advantage of the desirable properties of the NSME in this converter topology. FIG. 38 is a schematic for an isolated two-stage low side switch buck converter sub-circuit LSBKPP. This converter topology consists of a high efficiency low-side switch buck stage, which provides regulated power to an efficient push-pull isolation stage. An efficient center-tap fullwave rectifier provides rectification. Sub-circuit LSBKPP consists of diode D 46 , capacitor C 46 , FET transistor Q 141 , sub-circuit REG (FIG. 36 ), sub-circuit OUTB (FIG. 25A) sub-circuit BL 1 (FIG. 18 B), sub-circuit TCTP (FIG. 8 ), sub-circuit IFB (FIG. 40 B), sub-circuit AMP (FIG. 29 ), sub-circuit DCAC 1 , and sub-circuit PWFM (FIG. 33 ). FIG. 38 Table Element Value/part number C46 250 uf D46 MUR820 Q141 IRF540N External power source VBAT connects to pins DCIN+ and DCIN−. From pin DCIN+ connects to sub-circuit REG pin RIN+, D 46 cathode, capacitor C 46 , sub-circuit TCTP (FIG. 8) pin DCIN+, and sub-circuit DCAC 1 pin DC+. Voltage regulator sub-circuit REG output pin + 18 V connects to sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit REG provides regulated low voltage power to the controller and to the main switch buffer. VBAT negative is connected to pin DCIN− and ground return node GND. Node GND connects to sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , Q 141 source, sub-circuit IFB pin FBE, sub-circuit REG pin REG 0 and sub-circuit TCTP pin DCIN−. Sub-circuit PWFM (FIG. 33) is designed for variable pulse width operation. The nominal frequency is between 20-600 Khz PWFM is configured for maximum pulse width 90% (maximum buck voltage) with no feedback current from sub-circuit IFB. Increasing the feedback current reduces the Q 111 on time reducing the voltage to the push-pull stage and the output from converter LSBKPP. Sub-circuit PWFM clock output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP (FIG. 29 ). The output of switch speed up buffer sub-circuit AMP pin GA 2 is connected to the gate of Q 141 . Floating isolated 18-volt power from sub-circuit TCTP pin B+ connects to sub-circuit DCAC 1 pin P 18 V. The drain of Q 141 is connected to sub-circuit BL 1 pin P 1 A and the anode of D 46 . The return line of sub-circuit DCAC 1 pin DC− connects to sub-circuit BL 1 pin P 1 B, sub-circuit TCTP pin B− and C 46 . With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 141 . Switch Q 141 conducts reverse biasing diode D 46 ; capacitor C 46 starts charging through NSME BL 1 from source VBAT. During the time Q 141 is conducting, energy is stored in NSME sub-circuit BL 1 . Charging C 46 provides power to final push-pull converter stage DCAC 1 . The output of the output rectifier sub-circuit OUTB is connected to feedback sub-circuit IFB output pin FBC from sub-circuit IFB is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . Sub-circuit IFB removes current from PW 1 commanding PWFM to reduce the pulse-width or on time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PFFM switches CLK low turning off Q 141 stopping the current into BL 1 . The energy is released from NSME BL 1 into the now forward biased flyback diode D 46 charging capacitor C 46 . By modulating the on time of switch Q 141 the converter voltage is regulated. Regulated voltage is developed across C 46 Nodes DC+ and GND. Providing energy to the isolated constant frequency push-pull DC to AC converter sub-circuit DCAC 1 (FIG. 2 ). Sub-circuit DCAC 1 provides efficient conversion of the regulated buck voltage to a higher or lower voltage set by the magnetic element winding sub-circuit PPT 1 (FIG. 19) ratio. The center tap of the push-pull output magnetic is connected to, sub-circuit OUTB pin OUT−, RLOAD, sub-circuit IFB pin OUT− and the pin OUT− forming the return line for the load and feedback network. Output of sub-circuit DCAC 1 pin ACH is connected to sub-circuit OUTB pin C 7 B. Output of sub-circuit DCAC 1 pin ACL is connected to sub-circuit OUTB pin C 8 B. Sub-circuit OUTB provides rectification of the AC power generated by sub-circuit DCAC 1 . As the non-saturation magnetic converter is very quite minimal filtering is required by OUTB. This further reduces cost and improves efficiency as losses to filter components are minimized. Sub-circuit IFB provides the isolated feedback current to the sub-circuit PWFM. When sub-circuit IFB senses the converter output (nodes OUT+ and OUT−) is greater than the designed/desired voltage, current is removed from node PM 1 . Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus increasing the buck action and reducing the first stage converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 141 is removed stopping all buck activity capacitor discharging C 46 . Input current from VBAT to charge C 46 is sinusoidal making the converter very quiet. In addition the switch Q 141 is not exposed a potentially destructive current spike. Placing less stress on the switches thereby increasing the MTBF. Sub-circuit LSBKPP takes advantage of the desirable properties of the NSME in this converter topology. Adjusting the NSME BL 1 (FIG. 18B) sets the amount of buck voltage available to the final push-pull isolation stage. Greater efficiencies are achieved at higher voltages. The final output voltage is set by the turns ratio of the push-pull element PPT 1 (FIG. 19 ). Converter LSBKPP provides efficient conversion from high voltage sources into high current isolated output. FIG. 39 is a schematic for an isolated two-stage low side switch buck converter sub-circuit LSBKPPBR. This converter topology consists of a non-isolated high efficiency low-side switch buck stage, which provides regulated power to an efficient push-pull isolation stage. A fullwave bridge rectifier provides rectification. Sub-circuit LSBKPPBR consists of diode D 6 , capacitor C 6 , FET transistor Q 111 , sub-circuit REG (FIG. 36 ), sub-circuit OUTBB (FIG. 25 B), sub-circuit BL 1 (FIG. 18 B), sub-circuit TCTP (FIG. 8 ), sub-circuit IFB (FIG. 40 B), sub-circuit AMP (FIG. 29 ), sub-circuit DCAC 1 (FIG. 2 ), and sub-circuit PWFM (FIG. 33 ). FIG. 39 Table Element Value/part number C6 250 uf D6 MUR820 Q111 IRFP External power source VBAT connects to pins DCIN+ and DCIN−. From pin DCIN+ connects to sub-circuit REG pin RIN+, D 6 cathode, capacitor C 6 , sub-circuit TCTP (FIG. 8) pin DCIN+, and sub-circuit DCAC 1 pin DC+. Voltage regulator sub-circuit REG output pin + 18 V connects to sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit REG provides regulated low voltage power to the controller and to the main switch buffer. VBAT negative is connected to pin DCIN− connects to sub-circuit PWFM pin PWFM 0 , sub-circuit AMP pin GA 0 , Q 111 source, sub-circuit IFB pin FBE, sub-circuit REG pin REG 0 , sub-circuit TCTP pin DCIN−. Sub-circuit PWFM (FIG. 33) is designed for variable pulse width operation. The nominal frequency is between 20-600 Khz PWFM is configured for maximum pulse width 90% (maximum buck voltage) with no feedback current from sub-circuit IFB. Increasing the feedback current reduces the Q 111 on time reducing the voltage to the push-pull stage and the output from converter LSBKPPBR. Sub-circuit PWFM clock output pin CLK is connected to the input pin GA 1 of buffer sub-circuit AMP (FIG. 29 ). The output of switch speed up buffer sub-circuit AMP pin GA 2 is connected to the gate of Q 111 . Floating isolated 18-volt power from sub-circuit TCTP pin B+ connects to sub-circuit DCAC 1 pin P 18 V. The drain of Q 111 is connected to sub-circuit BL 1 pin PA 1 and the anode of D 6 . The return line of sub-circuit DCAC 1 pin DC-connects to sub-circuit BL 1 pin P 1 B, sub-circuit TCTP pin B− and C 6 . With sub-circuit PWFM pin CLK high buffer AMP output pin GA 2 charges the gate of transistor switch Q 111 . Switch Q 111 conducts reverse biasing diode D 6 ; capacitor C 6 starts charging through NSME BL 1 from source VBAT. During the time Q 111 is conducting, energy is stored in NSME sub-circuit BL 1 . Charging C 6 provides power to final push-pull converter stage DCAC 1 . The output of the output rectifier sub-circuit OUTBB is connected to feedback sub-circuit IPB output pin FBC from sub-circuit IFB is connected to sub-circuit PWFM pulse-width adjustment pin PW 1 . Sub-circuit IFB removes current from PW 1 commanding PWFM to reduce the pulse-width or on time of signal CLK. After sub-circuit PWFM reaches the commanded pulse-width PFFM switches CLK low turning off Q 111 stopping the current into BL 1 . The energy is released from NSME BL 1 into the now forward biased flyback diode D 6 charging capacitor C 6 . By modulating the on time of switch Q 111 the converter voltage is regulated. Regulated voltage is developed across C 6 nodes DC+ and DC−. Providing energy to the isolated constant frequency push-pull DC to AC converter sub-circuit DCAC 1 (FIG. 2 ). Sub-circuit DCAC 1 provides efficient conversion of the regulated buck voltage to a higher or lower voltage set by the magnetic element winding sub-circuit PPT 1 (FIG. 19) ratio. The return node of the sub-circuit OUTBB pin OUT− is connected to RLOAD, sub-circuit DCAC 1 pin AC 0 , sub-circuit IFB pin OUT− and the pin OUT−. Node OUT− is the return line for the load and feedback network. Output of sub-circuit DCAC 1 pin ACH is connected to sub-circuit OUTBB pin C 7 B. Output of sub-circuit DCAC 1 pin ACL is connected to sub-circuit OUTBB pin C 8 B. Sub-circuit OUTBB provides rectification of the AC power generated by sub-circuit DCAC 1 . As the disclosed non-saturation magnetic converter has minimal output ripple, less filtering is required by OUTBB. This further reduces cost and improves efficiency as losses in filter components are minimized. Sub-circuit IFB provides the isolated feedback current to the sub-circuit PWFM. Open collector output of IFB pin FBC connects to PWFM pin PW 1 . When sub-circuit IFB senses the converter output (nodes OUT+ and OUT−) is greater than the designed/desired voltage, current is removed from node PM 1 . Sinking current from PM 1 commands the PWFM to a shorter pulse-width thus increasing the buck action and reducing the first stage converter output voltage. In the event the feedback signal from IFB commands the PWFM to minimum output. Gate drive to switch Q 111 is removed stopping all buck activity capacitor discharging C 6 . As the NSME does not saturate the destructive noisy current spikes common to prior art are absent. Input current from VBAT to charge C 6 is sinusoidal making the converter very quiet. In addition the switch Q 111 is not exposed a potentially destructive current spike. Placing less stress on the switches thereby increasing the MTBF. Sub-circuit LSBKPPBR takes advantage of the desirable properties of the NSME in this converter topology. Adjusting the NSME BL 1 (FIG. 18B) sets the amount of buck voltage available to the final push-pull isolation stage. Greater efficiencies are achieved at higher voltages. The final output voltage is set by the turns ratio of the push-pull element PPT 1 (FIG. 19 ). Converter LSBKPPBR provides efficient conversion from high voltage sources such as high power factor AC to DC converters such as sub-circuit ACDCPF (FIG. 4 ). FIG. 40 PFC Over Voltage Feed Back Sub-circuit IPFFB FIG. 40 is the schematic of the inventions isolated over voltage feed back network sub-circuit IPFFB. Sub-circuit IPFFB consists of Resistors R 926 , R 927 , R 928 , R 929 and R 930 , capacitor C 927 , zener diodes D 928 and D 903 , transistor Q 915 and opto-isolator U 903 . FIG. 40 Table Element Value/part number U903 NEC2501 Q915 FZT705CT D903 ML5248B (18 v) D928 1SMB5956BT3 (200 v) R926 20k ohms R927 10k ohms R928 10k ohms R929 10k ohms R930 20k ohms Node PF+ connects through resistor R 927 to cathode of D 903 and anode of opto-isolator U 903 . Cathode of diode D 903 is connected to pin PF+. Resistor R 928 is connected from anode of D 928 to base of Q 915 . Capacitor C 927 is connected in parallel with zener diode D 903 . Resistor R 928 limits maximum base current. Resistor R 929 is connected between base and emitter of Q 915 . Resistor R 929 is used to shunt excess zener leakage current from the base common in high voltage diodes. Two hundred-volt zener diode cathodes D 928 are connected to pin PF+. Anode of D 928 is connected to R 930 and R 928 . Resistor R 930 provides a path for leakage current from 200-volt zener D 928 . Resistor R 926 limits the maximum current to U 903 internal light emitting diode to about 10 ma. Resistor R 927 sets the maximum zener current at maximum boost voltage of approximately 200-volts to 20 ma. Transistor Q 915 is biased off when the voltage from node PF+ and PF− is less than the zener voltage of 200-volts. Transistor is in a cutoff or non-conducting state no current is injected to U 903 LED. The internal phototransistor is also in a non-conducting state. The attached external control sub-circuit is not commanded to change its output. With 200 volts or more applied to nodes PF+ and PF− reverse biased zener diode D 928 injects current into the base of Q 915 . Resistor R 927 , capacitor C 927 and diode D 903 provide 18-volts to the collector of Q 915 . Transistor Q 915 conducts current into U 903 LED injecting base current into the U 903 phototransistor. Modulating the LED current is reflected as variable impedance between FBC and FBE. This phototransistor may be connected as a variable current source or impedance. This sub-circuit senses excessive boost voltage and quickly feeds back to the control sub-circuit (See PFA (FIG. 23 ), PFB (FIG. 24) or (PWFM FIG. 33 )) automatically reducing the boost voltage. FIG. 40A is a schematic diagram of the non-isolated boost output voltage feed back sub-circuit FBA. Sub-circuit FBA consists of Resistors R 1120 , R 1121 , R 1122 , R 1123 and R 1124 . Table FIG. 40A Value/part Element number R1123 499k ohms R1124 499k ohms R1122 6.65k ohms R1121 499k ohms R1120 1 MEG ohms Input node PF+ connected to series resistor [R 1123 +R 1124 ] then to parallel resistors [R 20 ||R 21 ||R 22 ] to the return node BR−. Resistors R 1120 , R 1121 , R 1122 , R 1123 and R 1124 values are selected for a nominal input voltage of 385-volts and output feed back voltage of 3.85. (See oscillograph G 1 FIG. 34) Resistors R 1120 , R 1121 , R 1122 , R 1123 and R 1124 are shown in surface mount configuration but can be combined into two thru hole-resistors. Feedback output node PF 1 is connected to node PF 1 of sub-circuit PFA (FIG. 23) or PFB (FIG. 24 ). Return pin BR− is connected to BR− of PFA (FIG. 23) or PFB (FIG. 24 ). Nodes FBE and FBC it may also be connected between nodes FM 1 pin PWFM 0 or PW 1 pin PWFM 0 of control sub-circuit PWFM (FIG. 33 ). FIG. 40B Output Voltage Feed Back Sub-circuit IFB FIG. 40B is the schematic of the inventions isolated low voltage feed back network sub-circuit FBA. Sub-circuit IFB consists of Resistors R 900 , R 901 and R 902 , zener diode D 900 , Darlington transistor Q 900 and opto-isolator U 900 . FIG. 40B Table Element Value/part number U900 NEC2501 Q900 FZT705CT D900 IN5261BDICT R900  1k ohms R902  4k ohms R901 40k ohms Node OUT+ connects cathode of D 900 to R 901 . Anode of diode D 900 is connected to series resistor R 900 to base of Darlington transistor Q 900 . Resistor R 902 is connected from base to emitter to Q 900 . Resistor R 901 connects to anode of opto-isolator U 900 LED (light emitting diode) the cathode is connected to Q 900 collector. Emitter of Q 900 is the return current path and connects to pin/node OUT−. Resistor R 901 limits the maximum current to U 900 internal light emitting diode to 20 ma. Resistor R 902 shunts some of the zener leakage current from the base. Zener diode voltage selection sets the converter output voltage a typical value maybe 48-volts. The zener voltage is the final desired output minus two base emitter junction drops (1.4V). Once the OUT+ node reaches the zener voltage a small base current biases Q 900 into a conducting state turning“on” opto-isolator U 900 internal LED. Resistor R 900 limits the maximum base current to Q 900 . Resistors R 900 and R 901 are selected to bias Darlington transistor Q 900 collector current with nominal voltage across nodes OUT+ and OUT−. Change in voltage between OUT+ and OUT− modulates the opto-isolator U 900 LED current in turn changing the base current of U 900 internal photo transistor. Phototransistor emitter is node FBE collector is node FBC. Modulating the LED current is reflected as variable impedance between FBC and FBE. This phototransistor may be connected as a variable current source or impedance. When used with control sub-circuit PFA (FIG. 23 ), PFB (FIG. 24) or (PWFM FIG. 33) the phototransistor is connected as a current shunt. Higher voltage applied to OUT+ and OUT− nodes increases the feedback shunt current commanding the control sub-circuit (See PFA (FIG. 23) or PFB (FIG. 24) or PWFM (FIG. 33 )) to reduce the pulse-width or frequency. IFB accomplishes high speed feed back due to the very high gain of the Darlington transistor and the rapid response of the internal converter stage(s) active ripple reduction and excellent load regulation are achieved. FIG. 40C is the schematic of the alternate PFC isolated over voltage feed back network sub-circuit IOVFB. Sub-circuit IOVFB consists of resistors of R 917 , R 938 , R 939 and R 940 , diode D 911 , Darlington transistor Q 914 and opto-isolator U 905 . FIG. 40C Table Element Value/part number U905 NEC2501 Q914 FZT705CT R938 160k ohms R939  70k ohms D911 1N5261BOTCT R940  50k ohms R917  40k ohms The output of the PFC at pin PF+ is connected to R 917 then to collector of Q 914 . Resistor R 917 sets the maximum current to U 905 light emitting diode. Resistor R 938 is connected from return node PF+ to zener diode D 911 cathode and R 938 . Resistor R 939 is connected from return node PF− to zener diode D 911 cathode and R 938 . Anode of D 911 is connected to wiper arm of adjustable resistor R 940 . One leg of R 940 is connected to the base of transistor Q 914 the other to R 939 and U 905 LED anode and R 939 . Emitter of Q 914 is connected to anode of U 904 . Adjustable resistor R 940 sets the maximum or trip voltage before transistor Q 914 is biased on. Providing current to U 905 LED. Phototransistor emitter is node FBE collector is node FBC. Modulating the LED current is reflected as variable impedance between FBC and FBE. This phototransistor is normally connected as a shunt to force the control element to a minimum output. This sub-circuit senses the boost voltage and feeds back to the PFC. Where excessive boost voltage forces the PFC to automatically reduce the boost voltage. FIG. 41 Output Voltage Feed Back Sub-circuit FBI FIG. 41 is the schematic of the alternate low voltage feed back network sub-circuit FBI. Sub-circuit FBI consists of resistors R 81 , R 82 and R 83 , zener diode D 80 , NPN transistor Q 80 and capacitor C 80 . FIG. 41 Table Element Value/part number R81 1k ohms D80 Zener Voltage = (Desired Output-0.65 V) Q80 BCX70KCT C80 1000 pf R82 1k ohms R83 715k ohms Node OUT+ connects cathode of D 80 . Anode of diode D 80 is connected to through resistor R 83 to OUT− and resistor R 82 to base of transistor Q 80 . Capacitor C 80 is connected from base to pin OUT−. Capacitor C 80 bypasses high frequency to noise to OUT−. Resistor R 81 is connected from emitter of Q 80 to node OUT−. Resistor R 81 adds local negative feedback to reduces the effects of variation in transistor gain. Collector of Q 80 is connected to pin FBC. The return current node connects to pins FBE and OUT−. Resistor R 82 limits the maximum base current protecting Q 80 . Resistor R 83 shunts some of the zener leakage current from the base. Zener diode voltage selection sets the converter output voltage a typical value maybe 48-volts. The zener voltage is the final desired output minus one base emitter junction drop (0.65-Volts). When the OUT+ node reaches the nominal level reverse biased zener starts to conduct injecting a small base current into Q 80 . Biasing transistor into a conducting state. Change in voltage between OUT+ and OUT− modulates Q 80 collector current. During normal operation the zener diode is biased at it's knee thus small changes in voltage result in relatively large collector current changes. When sub-circuit FBI used with control sub-circuit PFA (FIG. 23 ), PFB (FIG. 24) or (FIG. 33) the transistor is connected as a current shunt. Higher voltage applied to OUT+ and OUT− nodes increases the feedback shunt current commanding the control sub-circuit (See PFA (FIG. 23) or PFB (FIG. 24) or PWFM (FIG. 33) to reduce the pulse-width or frequency. Sub-circuit FBI provides high-speed feedback and gain to ripple components. With the rapid response of the internal converter stage(s) active ripple reduction and excellent load regulation are achieved. FIG. 42 Over Voltage Protection Sub-circuit OVP 1 FIG. 42 is the schematic of the inventions over voltage protection embodiment sub-circuit OVP 1 . Sub-circuit OVPL consists of SCR (silicon controlled rectifier) SCR 1200 , resistor R 1200 , capacitor C 1200 and zener diodes D 1200 , D 1202 and D 1203 . FIG. 42 Table Element Value/part number SCR1200 MCR265-10 D1203 BZT03-C200 (200 V) D1202 BZT03-C200 (200 V) D1200 IN4753 (5.1 v) C1200 220 pf R1200 10,0k ohms Input pin PF+ is connected to cathode of zener diode D 1203 , anode of D 1203 is connected to series zener diodes [D 1202 +D 1200 ] then to gate of SCR 1200 . Noise attenuation network of [R 1200 ||C 1200 ] is connected from SCR SCR 1200 gate to the return node BR−. Diodes D 1102 and D 1103 are both 200-volt; D 1101 is a 5.1-volt type the sum of the zener voltages set the trip point of the OVP at 405-volts. Other trip voltages may be implemented by selecting other zener diode combinations. Capacitor C 1200 and R 1200 prevents leakage current and transients from accidentally tripping the OVP. In the event of very high AC line voltages or a component failure in a feed back loop (FIG. 40A, 40 B, 40 C or 40 ) The boost voltage may quickly rise increase to levels dangerous to the output switch or output storage capacitors. When the output boost voltage of the at node PF+ rises above 405 V, zener diodes D 1203 , D 1202 and D 1200 conduct a small current into the gate of SCR 1200 turning SCR 1200 on. Turning SCR 1200 on places a low impedance path across the AC line through the rectifier sub-circuit BR (FIG. 22 ). SCR 1200 and bridge rectifier diodes must be selected to withstand the short circuit currents that may exceed 100 amperes until the input fuse opens. Thus quickly limiting the boost output voltage to a safe level. This circuit should never operate under normal AC line voltages. By changing zener voltages this sub-circuit would also be suitable for use in the across the rectifier output to protect the load from an over voltage condition. Sub-circuits OVP 1 shuts down the converter with out opening the line fuse. Sub-circuit OVP may be used in combination with OVP 1 (FIG. 42A) as a fail-safe back up for critical loads. FIG. 42A Over Voltage Protection Sub-circuit OVP 1 FIG. 42A is the schematic of the inventions over voltage protection embodiment sub-circuit OVP 2 . Sub-circuit OVP 2 consists of SCRs (silicon controlled rectifier) SCR 1101 and SCR 1100 , resistors R 1101 and R 1102 , capacitors C 1100 and C 1101 and zener diodes D 1100 , D 1102 and D 1103 . FIG. 42A Table Element Value/part number SCR1101 S101E (Teccor) SCR1100 S601E (Teccor) D1103 BZT03-C200 (200 V) D1102 BZT03-C200 (200 V) D1100 IN4753 (5.1 v) R1100 16000 R1101 5.1K ohms R1102 5.1K ohms C1100 1200 pf C1101 1200 pf Anode of SCR 1101 is node/pin CP 18 V+ that is connected to external control DC source. Return node BR− is connected to SCR 1101 cathode and capacitor C 1100 . Input node PF+ is connected to cathode of zener diode D 1103 and to series resistor R 1100 then to anode of SCR SCR 1102 . The anode of D 1103 is connected to the cathode of D 1102 . The anode of 10 D 1102 is connected to the cathode of D 1100 . The cathode of SCR 1100 is connected to the gate of SCR 1101 . The anode of D 1103 is connected to series zener diodes [D 1102 +D 1100 ] then to capacitor C 1100 then to the return node BR−. Capacitor [C 1200 ||R 1200 ] prevents leakage current and transients from accidentally tripping OVPB. In the event of very high AC line voltages or a component failure in a feed back loop (IPFFB FIG. 40A, FBA 40 B, IFB 40 C or FBI FIG. 41) The boost voltage may quickly rise increase to levels dangerous to the output switch or output storage capacitors. When the output boost voltage of the at node PF+ rises above 405V, zener diodes D 1103 , D 1102 and D 1100 conduct a small current into the gate of SCR 1001 latching SCR 1001 on. Resistor R 1100 provides holding current for SCR 1101 . Turning SCR 1101 provides gate current to SCR 1100 , resistors R 1100 and R 1101 limits the gate current and provides the hold current to SCR 1100 . With gate current to SCR 1100 the SCR is turned on providing a low impedance path from nodes CP 18 V+ to BR−. This action removes the regulated power to the main switch buffer and or PWM controllers PFA (FIG. 23) or PWFM (FIG. 33) and or buffer AMP (FIG. 29) thus turning off the main switch. The converter is held in an off state until boost voltage PF+ through R 1100 can not maintain the holding current of SCR 101 . Typically power must be removed from the system to reset SCR 1101 . The minimum holding current of SCR 1001 is typically 5-10 ma. The action of OVP 1 quickly limits the boost output voltage to a safe level. This circuit should never operate under normal AC line voltages. By changing zener voltages this sub-circuit would also be suitable for use across the output rectifier to protect the load from an over voltage condition. Sub-circuit OVP 1 gracefully shuts down the converter requiring manual intervention to reset the fault. FIG. 42B is the schematic of the isolated output over voltage feed back network sub-circuit OVP 2 . Sub-circuit OVP 2 consists of resistors of R 970 , R 971 , and R 972 , capacitor C 970 , zener diode D 970 , SCR SCR 970 , Darlington transistor Q 970 and opto-isolator U 970 . FIG. 42B Table Element Value/part number D970 1N5261BOTCT U970 NEC2501 Q970 FZT705CT R970 160k ohms R971  10k ohms R972  22k ohms C970  200 pf The output of the converter at pin OUT+ is connected to R 972 and to the cathode of zener diode D 970 . The anode of D 970 is connected to series resistor R 970 then to base of Q 970 . Resistor R 970 sets the maximum base current to Q 970 . Resistor R 971 is connected between the anode of D 970 and return node OUT− Anode of light emitting diode U 970 is connected to resistor R 972 then to OUT+. The cathode of U 970 LED is connected to Q 980 collector. Emitter of Q 980 is connected to return node OUT−. Zener diode D 960 sets the maximum or trip voltage before transistor Q 970 is biased on providing current to U 970 LED. Application of voltage greater than the zener voltage of D 970 injects a small base current into Q 970 . Transistor Q 970 turns on the internal LED of U 970 placing phototransistor in a conducting state and low impedance to pins OVC and OVC. External push-pull driver sub-circuit PPG (FIG. 43) is shut down immediately by bringing pin PPEN high stopping the output stage. Sub-circuit OVP 2 senses the output voltage and quickly feeds back to the push-pull PFC. Where excessive boost voltage forces the PFC to automatically reduce the boost voltage. FIG. 42C is the schematic of the isolated output over voltage crowbar network sub-circuit OVP 3 . Sub-circuit OVP 3 consists of resistors of R 980 , R 981 , R 982 , R 983 , R 984 and R 985 , capacitors C 980 , C 981 and C 982 zener diode D 980 , SCRs SCR 980 and SCR 981 , Darlington transistor Q 980 and opto-isolator U 980 . FIG. 42C Table Element Value/part number D980 1N5261BOTCT SR980 S601E (Teccor) U980 NEC2501 Q980 FZT705CT R980 160k ohms R981 10k ohms R982 22k ohms R983 51k ohms R984 1200 ohms R985 510 ohms C980 200 pf C981 1200 pf C982 1200 pf The converter output is sensed at pin OUT+ reference to pin OUT−. Pin OUT+ is connected to resistor R 982 and to the cathode of zener diode D 980 . The anode of D 980 is connected to series resistor R 980 then to base of Q 980 . Resistor R 980 limits the base current to Q 980 . Resistor R 981 is connected between the anode of D 980 and return node OUT− to provide a diode leakage current path. Anode of light emitting diode U 980 is connected through resistor R 982 then to OUT+. The cathode of U 980 LED is connected to Q 980 collector. Emitter of Q 980 is connected to return node OUT−. Zener diode D 960 sets the maximum or trip voltage before transistor Q 980 is biased on providing current to U 980 LED. Application of voltage greater than the zener voltage of D 980 injects a small base current into Q 980 . Emitter of opto-isolator U 980 is connected to the gate of SCR 981 and through [R 984 ||C 982 ] to return node BR−. Transistor Q 980 turns on the internal LED of U 980 placing phototransistor in a conducting state and supplying gate current to SRC SCR 981 from the external 18-volt source connected to pin CP 18 V+. Network [R 984 ||C 982 ] prevents false triggering of SCR SCR 981 . The cathode of SCR SCR 981 is connected to the gate of SCR SCR 980 and through [R 985 ||C 981 ] to return BR−. With SCR SCR 981 turned on gate current is provided to low voltage SCR SCR 980 . High voltage boost output is connected to pin PF+ resistor R 983 supplies hold current to SCR SCR 981 holding SCR SCR 980 on. SCR SCR 980 is selected for low hold current and ability to block the maximum boost voltage on PF+. SCR SRC 980 anode is connected to pin CP 18 V+. SCR SRC 980 cathode is connected to return pin BR−. SCR 980 clamps the low voltage supply CP (FIG. 26) or CPA (FIG. 27 ). With the low supply held down the gate drive to the main switch is disabled turning off the converter. With the main switch Q 1 (FIG. 1, 3 , 4 ) turned off holdup capacitor C 17 charges to applied AC line peak. With pin PF+ held near line peak SCRs SCR 981 will hold SCR SCR 981 on until AC line power is removed to the converter. Sub-circuit OVP 3 senses the out of specification output voltage and quickly stop the converter thereby protecting the load and converter with out generating destructive currents like OVP (FIG. 42 ). FIG. 43 Push-pull oscillator sub-circuit PPG FIG. 43 is the push-pull oscillator sub-circuit of the invention. The current implementation uses a Motorola MC33025 pulse width modulator IC to generate the clock signals to drive the push-pull output stage. Sub-circuit PPG consists of U 14 a two-phase oscillator, resistors R 126 , R 130 , R 131 , R 132 , R 133 , R 134 , R 135 , R 136 and R 137 , capacitors C 143 , C 136 , C 139 , C 140 , C 141 and C 142 . FIG. 43 Table Element Value/part number U14 MC33025 R126 12k ohms R130 10 ohms R131 10 ohms R132 47k ohms R133 10k ohms R134 100k ohms R135 15k ohms R136 1.5 MEG ohms R137 15k ohms C136 0.22 uf C139 0.22 uf C140 0.22 uf C141 0.01 uf C142 0.001 uf C143 .33 uf The current implementation uses a Motorola MC33025 pulse width modulator IC to generate the clock signals to drive the push-pull stage. But, any non-overlapping two phase fixed frequency generator could be used. Pin 1 of U 14 is connected to [capacitor C 143 ||Resistor R 132 ] then to pin 3 . Resistor R 134 connects the internal 5.1-volt reference output of U 14 pin 16 to pin 1 . Resistors R 135 in series with R 137 from 5.1-volt reference to return node PPG 0 form a voltage divider; the center is connected to U 14 pin 2 placing pin 2 at 2.55-volts. Resistor R 126 is connected from U 14 pin 5 to return node PPG 0 . Resistor R 133 is connected from U 14 pin 1 to return node PPG 0 . Timing capacitor C 142 is connected from U 14 pins 6 and 7 to return node PPG 0 . Resistor R 126 and capacitor C 142 set the operating frequency of the internal oscillator. Timing resistor could be replaced with a JFET, MOSFET, transistor, or similar switching device to provide variable frequency operation. The drain of the transistor would be connected to pin 5 . The source would be connected to return node PPG 0 . The variable frequency command voltage/current is applied between gate and source. Capacitor C 141 is connected from U 14 pin 8 to return node PPG 0 . Capacitor C 136 is connected from U 14 pin 16 to return node PPG 0 . Capacitor C 140 is connected from U 14 pin 15 to return node PPG 0 . Capacitor C 139 is connected from U 14 pin 13 to return node PPG 0 . Resistor R 136 is connected from U 14 pin 9 to return node PPG 0 . U 14 pins 10 and 12 is connected to return node PPG 0 . External power is connected to node/pin PPG+ to PWM (pulse width modulator) IC U 14 on pin 15 through resistor R 130 connected to the 18-volt control supply. Resistor R 131 connected to pin 13 of U 14 and PPG+ provides power to the totem-poll output stage. The power return line is connected to node PPG 0 . IC U 14 is designed to operate at a constant frequency of approximately 20-600 Khz with a fixed duty cycle of 35-49.9%. Resistors R 135 , R 137 , R 133 configure U 14 to operate at maximum pulse width. A two-phase non-over lapping square wave is generated on pins 11 node PH 2 and pin 14 node PH 1 and delivered to speed-up buffers AMP described in FIG. 29 . The two-phase generator is configured to prevent the issue of overlapping drive signals that would null the core bias and present excessive current to the switches. Sub-circuit PPG provides the drive to the push-pull switches making efficient use of the NSME. Although the present invention has been described with reference to a preferred embodiment, numerous modifications and variations can be made and still the result will come within the scope of the invention. No limitation with respect to the specific embodiments disclosed herein is intended or should be inferred.

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